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A 2kW 24V/240VAC sinewave inverter; Pt.2 This month, we present the DC to DC converter circuitry of our 2kW sinewave inverter. This circuit steps up from 24V DC to 365V DC & uses Mosfets to drive a high-frequency transformer. By JOHN CLARKE Converting 24V DC to 365V DC at a power level of more th an ZkW is n o simple task. If the step-up transformer is to be kept reasonably small an d light in w eight, it must run at qu ite a high frequ ency an d this presents considerable problems in making sure that the Mosfets are switched cleanly. Special measures are required. to ensure that interaction between the con- trol an d switching circuitry is kept to an absolute minimum. Befo re we dive into the full circuit description, let's have a look at a basic converter circuit so that we can get a picture of what h appens. Fig.5 shows the sch ematic of a basic converter. It uses a centre-tapped step-up transfo rmer which is driven by two Mosfets . The secondary winding of the transformer is fed to a bridge rectifier and a capacitor (Cl) to develop the DC output. The Mosfets are driven by a pulse width modulator which has feedback applied from the DC output via a voltage divider consisting of Rl and RZ. This feedback acts to reduce the width of the pulses from the pulse width modulator if the voltage starts to rise. Similarly, if the voltage starts to drop , the width of the pulses is increased. Push-pull drive Note that the centre-tap of the transformer's primary winding is connected to +24V DC, while the two legs of the primary winding are each connected to a Mosfet. The pulse width modula- tor drives the gates of the Mosfets with a complementary square w ave signal so that when Mosfet Ql is on, Q2 is off and when Ql is off, Q2 is on. Now consider what happens when Ql is on and Q2 is off. With Ql on, the full 24V DC is applied to the top half of the transformer primary winding; ie, Ql 's drain is virtually at zero volts. At the very instant that current starts to flow in the top half of the transformer winding, transformer action operates so that 24V appears across the other half of the transformer primary winding. This means that when Ql is on, the drain of QZ will have +48V applied to it. Similarly, when Q2 turns on and Ql turns off, QZ 's drain will be at OV and Ql 's drain will be at +48V. So if you can imagine a complementary square wave applied to the Mosfet gates, then the primary of the transformer will have a square wave of 48V peak-to-peak applied to it, or 24V RMS. This is stepped up by the transformer so that the secondary voltage will be 24V multiplied by the turns ratio - in this case, 18:1. The secondary voltage therefore will be a square wave of 864V peak-to-peak or 432V RMS. This AC waveform is rectified by the bridge rectifier and filtered by capacitor Cl to give 432V DC. Note that, RE CTIFIER PULSE WIDTH MODULATOR Q2 + o--------' 24V '------------+--0- Fig.5: this block diagram shows the basic converter circuit. It uses a pulse width modulator to drive complementary Mosfet stages & these in turn drive a centretapped step-up transformer. The transformer output is then fed to a bridge rectifier & to capacitor Cl to produce 365V DC. for the purposes of th is description , we have neglected any losses which occur in the Mosfets, the transformer, the rectifier or filter capacitor. If you now look closely at Fig.5, you will notice that its output is shown as 365V. Why does this differ from the figure of 432V just mentioned? Well, remember the fe edback referred to earlier? It controls the width of the pulses applied to the Mosfet gates . So the Mosfets don 't actually have a full square wave app lied to them. Instead, the pulses fed to the gates have a duty cycle that's less th an 50% (ie, they're on for less th an 50% of th e time). This means that the circuit has plenty of margin so that it can maintain the output at 365V, regardless of variations in the load an d input voltage. The full circuit Fig.6 shows the full circuit details. It uses 24 transistors, 23 ICs , 13 di- Below: this close-up view shows the DC to DC converter board, with part of the high-frequency transformer visible in the foreground. The 12 Mosfet transistors are bolted to the chassis on either side of the PC board. NOVEMBE R 1992 25 :;; n:i: z n 0 r=: CFJ N Cl PO~R L....o--:"'1 0 T OF~Ao MOTOR START 4.7k f uoo 151 IN2 1r-e 3 1l•IN1 116 ~a ; .001! +IN2 E1 1g 10k 10k f f 150k +15V ,.. f T 101 .Ii:' 35VWJ f 101 ..:- IOUT 35VWl 01t .,. E21101 I 5 112 vcc _t;_T IC1 TL494 111 C2 - T +15V ~w V#t +24V DC-DC CONVERTER j 27k : 41DEAD TIME RT ' 141VAEF 13 ~ E'...:___:JC! VIEWED FROM' BELOW• r-'\ -e- * • ♦ + r-11 ' f .,. 10 25VW ,_ 47k ; ♦ t I o.1 I T A POWER • LED1 • 18 UNDER VOLTAGE DETECT 21 Cl - IN1 . .,. 10k ~ L1 100uH 1A ...L. ~SQ5-Q16 S !:I:- 24V : F1 100A 125V -0--.0-- I +24V SWITCHED 47k .,. 385V ,- •OUTPUTI: 01I ADJUST VR1 .,. 100k -.,. V-, I CHASSIS 400VD1 0.1· IC5 IL300 i.73 4.7M (1J~37) ~ 10k 0.5W T .......,.___. D3-D6 4xMUR1560 I 561/ .0011 •12k ZD6 ( j + l 15V 1W ·0.1 r n n ~~- -----------~~----~ --- ------~ r,-----,-~) ~-- +GND ., +15V ., 680 400VW, , _ OV F4 15AI +365V OUTPUT CONTROL VCC 13 OSCILLATOR Q t--,:J-+-t FLIPFLOP CK Q~ ~ - t D RT .,. 11 10 .,. 12 VCC .,. UV LOCKOUT ERROR AMP 1 REFERENCE REGULATOR o--+--J 16 FEEDBACK PWM COMPARATOR INPUT ERROR AMP 2 15 14 REF OUTPUT Fig. 7: this block diagram shows the internal circuitry of the TL494 PWM controller. It includes a sawtooth oscillator, two error amplifiers, a PWM comparator, a dead-time control comparator & a 5V reference. Emitter followers Ql & Q2 provide the complementary output signals at pins 9 & 10. odes and 14 zener diodes. There are also several inductors and transformers, plus numerous capacitors and resistors. Many of the components are specially selected for this application, while others are relatively common. At the heart of the DC-DC converter is a dedicated switchmode integrated circuit, a TL494 from Texas Instruments. It contains all the necessary circuitry to generate complementary square wave pulses at pins 9 & 10, to drive the gates of the switching Mosfets. Facilities for output voltage regulation and under voltage dropout are provided. The TL494 is equivalen l Lu Lhe "pulse width modulator" shown in the basic circuit ofFig.5. Paralleled Mosfets While the simple circuit of Fig.5 Fig.6 (left): the full circuit for the DCDC converter. ICl is the pulse width modulation controller. It generates complementary square wave outputs at pins 9 & 10 and these drive the Mosfet switching devices (Q5-Q10 on one side & Q11-Q16 on the other) via buffer stages IC2a-IC2f & transistors Q1-Q4. IC4, IC5 & IC3 provide voltage feedback to control the pulse width modulator. shows just two Mosfets driving the step-up circuit, the real circuit uses 12 Mosfets (six per side). These drive paralleled windings of one turn on the transformer, in order to carry the very large currents required (in excess of 100A at full load). Just how you make a 1-turn winding which carries these huge currents is a story in itself and we'll give those details in a future issue. On the secondary side of the transformer, four bifilar windings drive a bridge rectifier made up of soft recovery diodes D3-D6. The output from this bridge rectifier then charges a 680µF 400VW electrolytic capacitor via a lO0µH toroidal inductor. This inductor is there to reduce the peak charging currents through the diodes in the bridge rectifier. The voltage feedback system runs along the bottom of Fig.6. The output voltage (ie, across the 680µF capacitor) is sampled by a voltage divider (4.7MQ & 12kQ) and applied to pin 5 ofIC4 which then drives optocoupler IC5. IC5 then drives op amp IC3 which in turn drives an internal error amplifier in IC1 to control the pulse width modulation. The complementary outputs from IC1 appear at pins 9 & 10 and drive paralleled Mosfets Q5-Q16 via CMOS buffer stages and bipolar transistors GND Q1-Q4. These stages provide the necessary current amplification to drive the gates of the Mosfet stages. The TL494 So that's the converter circuit description in a nutshell. Now let's look at it in more detail, starting with IC1, the TL494. A block diagram showing the internal features of this device is shown in Fig. 7. It is a fixed frequency pulse width modulation (PWM) controller containing a sawtooth oscillator, two error amplifiers and a PWM comparator. It also includes a dead-time control comparator, a 5V reference and output control options for push-pull or single ended operation. The PWM comparator generates variable width output pulses by comparing the sawtooth oscillator waveforms with the outputs of the two error amplifiers. By virtue of the di ode gating system, the error amplifier with the highest output sets the pulse width. Fig.9 shows the two output waveforms generated by IC1. Dead~time comparator The dead time comparator prevents the push-pull outputs at pins 9 & 10 from changing over at the same time. It does this by providing a brief delay between one output swinging low and the other swinging high (ie , both outputs are low for a short time at the transition points). This delay is called NOVEMBER 1992 27 PARTS LIST FOR THE 2kW 24V/240VAC SINEWAVE INVERTER 1 aluminium case, 400 x 400 x 170mm 4 170mm-long radial finned heatsinks 8 4mm-dia. screws & nuts for securing heatsink 1 self-adhesive front panel label, 230 x 170mm 2 25mm cable entry glands (Clipsal 282/25) 1 Delta 10DRCG5 mains filter 1 dual power point, 250V 10A 2 2AG 250VAC panel mount fuse holders 1 500mA 2AG fuse (F2) 1 10A 2AG fuse (F5) 1 5mm LED & bezel (LED 1) 1 panel mount rocker switch (S1) 2 80°C thermal cutouts (TH 1, TH2) 8 large rubber feet 16 6mm metal standoffs plus screws & nuts 12 12mm metal standoffs plus screws & nuts 2 6. 7mm nylon cable clamps 27 100mm cable ties 4 200mm cable ties 6 150A tinned copper cable lugs with 10mm mounting holes 47 insulated crimp lugs, 4mm stud size, 2.5mm wire size 9 insulated crimp lugs, female quick connect, 2.5mm wire size 2 solder lugs 1 50ml packet of heatsink compound Wire & cable 1 1.5m-length 130A black automotive cable , 19610.4mm (25mm 2) 1 1.5m-length 130A red automotive cable, 19610.4mm (25mm 2 ) 1 Sm-length 30A 240VAC cable, 4110.32mm (3.3mm 2) 1 2.5m -length red or brown 15A 240V mains cable 1 2.5m-length black or blue 15A 240V mains cable "dead time" and constitutes about 5% of the switching time available. Dead time is essential in a very high power push-pull circuit such as this. Without it, the Mosfets driving one half of the step-up transformer 28 SrucoN CrnP 1 50mm -length green/yellow 15A 240V mains cable 1 2.'?m-length yellow heavy duty hook-up wire 1 2m-length green heavy duty hook-up wire 1 3m-length blue heavy duty hookup wire 1 2.2m 0 1ength red heavy duty hookup wire 1 500mm-length black heavy duty hook-up wire 1 1m-length 2-core shielded cable Primary PC board 1 PC board, code SC 1130992·1 , 204 x 157mm 9 PC stakes 12 4mm brass nuts, screws & shakeproof washers 1 200mm-length 0.8mm tinned copper wire 1 100µH 1A choke (Siemens B82111-E-C25) (L1) 1 3mm screw & nut for 3-terminal regulator 12 TOP-3 mica washers, bushes, 15mm long screws & nuts 1 100kQ horizontal mount cermet trimmer (VR1) Semiconductors 1 TL494 switchmode controller (IC1) 1 4050 hex buffer (IC2) 1 LM358 dual op amp (IC3) 2 BC338 NPN transistors (01, 03) 2 BC328 PNP transistors (02, 04) 12 Siemens BUZ349 N-Channel 30A 100V Mosfets (05-016) 1 7815 3-terminal regulator (REG1) 1 33V 1W zener diode (ZD1) 2 75V 1W zener diodes (ZD2, ZD3) 2 18V 1W zener diodes (ZD4, ZD5) 2 1N4002 1A diodes (D 1, D2) 4 0.1 µF 63VW MKT polyester 1 0.01 µF 63VW MKT polyester Resistors (0.25W, 1%) 3 1Mn 1 3 .3kQ 1 150kQ 1 2.2kQ 0.5W 3 47kQ 1 82!.15W 1 27kQ 12 10!.1 3 10kQ 2 4.7Q 2 4.7kQ 24 1Q Transformer board 1 fibreglass sheet, 175 x 370mm (eg, PC board material) 2 U93/76/30 N27 cores (Siemens B67345-B1 -X27) 2 coil formers (Siemens B67345A 1000sT1) 1 sheet of aluminium, 135 x 76 x 1.6mm 2 8 x 80mm bolts plus nuts & washers 1 100A/125A motor start cartridge fuse, lug mounting, 94mm centres (Hawker Siddeley CEO100M125, GEC TCP100M125) (F1) 2 Clipsal 2DLA 12C 165A brass link bars 2 Clipsal BP165C12 165A brass link bars 2 Clipsal DLA6 165A brass link bars 14 screws for mounting link bars 2 4-way 30A PVC 240VAC mains connector strips (Clipsal BP535 or 593/30) 2 3mm countersunk screws & nuts for mounting connector strip 2 3mm screws & nuts for mounting connector strip 4 pieces of 28 x 60mm 0.6mm sheet copper 4 solder lugs Capacitors Capacitors 3 10µF 35VW PC electrolytic 2 0.47µF 63VW MKT polyester would still be switching off while the Mosfets driving the other half of the transformer were switching on. This would place a brief but direct short circuit across the 24V battery supply. As a result, the Mosfets would be 8 10µF 63VW MKT (Siemens B32523-B106-K) 1 0.1 µF 400VW polyester capacitor destroyed - they would literally blow them themselves apart. Under-voltage cutout In Fig.6 , one of the error amplifiers in IC1 is used to provide the under Rectifier PC board 1 PC board, code SC11309924, 214x162mm 1 100 x 55mm sheet of fibreglass material 2 45 x 45mm sheets of fibreglass material 2 20 x 20 x 12mm aluminium right angle brackets (1 .6mm thick) 2 3 x 10mm screws & nuts for brackets 1 8 x 80mm bolt & nut 1 6mm ID rubber grommet · 1 45mm-dia. vertical mount capacitor clamp 2 3mm screws & nuts for clamp 2 0.75mH 10A toroid chokes (L3, L4) 1 iron powdered ring core , Neosid 17-745-22 (L2) 1 500mm-length of 2mm enamel led copper wire 1 50mm -length 0.8mm tinned copper wire 2 PC stakes 4 3AG fuse clips 2 15A 3AG fuses (F3, F4) 1 4-way 30A 240VAC mains connector strip 2 $ x 10mm screws & nuts for connector strip 8 4mm brass nuts plus star washers & screws · 4 TO220 mica washers , insulating bushes, screws & nuts Semiconductors 1 LM358 dual op amp (IC4) 1 Siemens IL300 linear optocoupler (IC5) 4 MUR1560 15A fast recovery 600V diodes (D3-D6) 1 15V 1W zener diode (ZD6) Capacitors 1 680µF 400VW LL electrolytic (Siemens B43570-E0687-O) 1 25µF 370VAC motor start capacitor with spade lug connectors (Plessey P331 1340) 1 0.1 µF 63VW MKT polyester 1 .001 µF 63VW MKT polyester voltage cutout feature. This prevents the batteries fro m being excessively discharged . Pin 2, the inverting input, is connected to th e 24V supply via a voltage divider consisting of a lOkQ resistor and a 3.3kQ resistor. Resistors (0.25W, 1%) 1 4.7MQ high voltage resistor (Philips VR37) 1 56kQ 510kQ 1W 1 12kQ 1 820Q Resistors (0.25W, 1%) 4 4.7kQ 410Q 2 100Q 41 Q 1 47Q Sinewave PC board H-drive PC board 1 PC board, code SC11309922, 204 x 157mm 1 50mm-length insulating sleeving 4 4mm brass nuts, screws & star washers 14 PC stakes 1 300mm-length 0.8mm tinned copper wire 8 metal oxide varistors (Siemens S14K275) 16 TOP-3 mica washers, insulating bushes , 12mm screws & nuts 3 EFD15/8/5 ferrite transformers (T2, T3 , T4) (ie, each with 2 x Philips 4312 020 41001 cores, 1 x Philips 4322 021 35201 former & 2 x Philips 4322 121 35141 clips) 1 2.5m-length of 0.125mm enamelled copper wire 1 200mm length of insulation tape Semiconductors 4 Siemens SFH6136 fast optocouplers (IC6, IC8, IC10, IC12) 4 4049 CMOS hex inverters (IC?, IC9, IC11 , IC13) 4 Siemens BUP304 1000V 35A IGBTs (017-020) 3 15V 1W zener diodes (ZD7, ZD9, ZD11) 4 18V 1W zener diodes (ZD8, ZD10, ZD12, Z013) 3 1N4148, 1N914 switching diodes (D7-D9) 4 Siemens BYP102 1000V 50A diodes (010-013) Capacitors 3 10µF 25VW PC electrolytics 4 1µF 50VW RBLL (low leakage) PC electrolytics 1 1µF 500VW axial electrolytic (Siemens B25839~B6105-K) Pin 1, th e non-inverting in put , is conn ected the chip's 5V internal referen ce (Vref) at pin 14 , via a 4.7kQ resistor. When the voltage at pin 2 drops below 5V (ie, when the battery volt- 1 PC board , code SC11309923, 181 x 131mm 17 PC stakes 1 700mm-length 0.8mm tinned copper wire 1 miniature TO-220 heatsink (Thermaloy 6038 type) 1 3.2768MHz parallel resonant crystal , 22pF loading (X1) 1 3 x 10mm screw & nut Semiconductors 1 7555 CMOS timer (IC1 4) 2 4049 hex inverters (IC15, IC16) 1 74HC04 hex inverter (IC17) 4 74HC193 4-bit binary counters (IC18-IC21) 1 74HC08 quad 2-input AND gate (IC22) 1 NMC27C64N250 CMOS OTP PROM (IC23) 4 BC337 NPN transistors (021-024) 1 7805 3-terminal regulator (REG2) 1 33V 1W zener diode (ZD14) Capacitors 3 10µF 35VW PC electrolytic 11 0.1 µF 63VW MKT polyester 1 220pF 63VW MKT polyester 2 22pF ceramic Resistors (0.25W, 1%) 1 10MQ 1 2.2kQ 1 100kQ 4 220Q 1 47kQ 1 150Q 5W 4 2.7kQ Note: this project has been sponsored by Rod Irving Electronics (1992) & full kits will be available from this company in ea rly 1993. Copyright of th e associated PC boards is assigned to Rod Irving Electronics. age is below 20V), the output of this error amp li fier goes high (pin 3) and switch es off both outputs at p ins 9 & 10. This effectively shuts down th e circuit. Note that the voltage divider fee dNovEMBER 1992 29 Fig.8: the inverter delivers a clean sinewave, as these scope photos show. It provides 250VAC at no load (top), · 243VAC at lkW (centre) & 230VAC at 2kW (bottom). Note that the waveform improves with increasing load. ing pin 2 is connected via two thermal cutout switches. These are mounted on two of the four heatsinks in the inverter chassis and are preset to open when the temperature exceeds 80°C. Thus, when one or both of the heatsinks becomes too hot, the thermal cutout opens and the circuit is shut down. When the heatsinks cool down, normal operation resumes. Voltage feedback The second error amplifier in !Cl is used to control the output voltage of the converter. The feedback voltage from the optocoupler and IC3 is fed to pin 16 and compared to the internal 30 SILICON CHIP 5V reference, applied to the pin 15 input via a 4. 7k0 resistor. Normally, the feedback voltage from IC3 should be close to 5V. If the output rises above this, the output of the error amplifier also rises and this reduces. the output pulse width. Conversely, if the output voltage falls, the error amplifier output also falls and the pulse width increases. The gain of this error amplifier at low frequencies is set by the lMO feedback resistor between pins 3 & 15 (giving a gain of 213). At higher frequencies, the gain is set is set to 22 by virtue of the 47k0 resistor and O. lµF capacitor in series across the lMO resistor. This reduction in gain at high frequencies prevents the error amplifier from responding to hash on the supply lines. The 27k0 resistor and .OOlµF capacitor at pins 6 and 5 of !Cl set the internal oscillator to about 40kHz. This is divided by the internal flipflop to give complementary output signals from pins 9 & 10 and so the resultant switching speed of the Mosfets is ZOkHz. Pin 4 of !Cl is the dead-time control. When this input is at the same level as Vref, the outputs at pins 9 & 10 are off. As pin 4 drops to OV, the dead-time decreases to a minimum. At initial switch on, the lOµF capacitor between Vref (pin 14) and pin 4 is discharged. This prevents the output transistors in !Cl from switching on. The lOµF capacitor then charges via the associated 47k0 resistor and so the duty cycle of the output transistors slowly increases until full control is gained by the error amplifier. This provides a soft start for the converter. Complementary outputs The outputs at pins 9 & 10 of !Cl come from internal emitter follower transistors and these each drive 10k0 load resistors. These outputs also each drive three paralleled CMOS non-inverting buffers (IC2a-IC2f). These buffers stages then drive transistors Ql & QZ on one side of the circuit, and Q3 & Q4 on the other. Thus, when pin 10 ofICl goes high, Ql turns on and drives the paralleled gates of Mosfets Q5-Q10 via a 4. 70 resistor. Note that each Mosfet gate is connected via a .100 "stopper" resistor to minimise any parasitic oscillations which may occur while the Fig.9: these are the complementary pulse signals from the TL494 PWM controller. Note that both waveforms do not switch over at the same points, in order to give dead time. Mosfets are switching on and off. When pin 10 goes low again, QZ switches on to discharge the gate capacitance of each Mosfet, thus switching them off. The complementary process occurs with pin 9. Circulating currents While it is not obvious from the circuit of Fig.6 , there must be two connections to the drain and source of each Mosfet. For example, the drains ofMosfets Q5-Q10 are connected in parallel to one side of the transformer primary. In addition, each drain must be connected back to the PC board so that the zener diode protection can work. Similarly, the source connections of the Mosfets are all connected to the OV line and thence to the battery negative pole. In addition, the sources all need to go back to th~ PC board because otherwise the gate drive circuitry would not work properly. The problem with this need for double connections is that unless we take precautions to stop it, very heavy currents will flow on the relatively flimsy copper tracks of the PC board. This must be avoided because the PC board cannot carry such currents. To stop heavy currents flowing on the board, 10 resistors are used in all the drain and source connections. Thus , the gate circuitry and zener diode protection circuitry works properly but the main currents flow in the direct cable connections. But even with this precaution, the source currents from the Mosfets still tend to flow in the PC board tracks. To stop this, two grounds are provided. The first is for the load current and connects directly to the negative ter- Despite the complex circuit, the 2kW Sinewave Inverter is relatively easy to build since most of the parts are mounted on PC boards. The DC-DC converter board is at the bottom of the photograph, with the high-frequency switching transformer immediately above it. minal of the battery. The second provides the circu it earth and is connected to the n egative battery terminal via a 100µH ch oke (11). This choke prevents the load current from flowing through the parallel H2 source resistors. It does this because the inductance between the PC board and the negative battery terminal is far greater than the in ductance of the source leads. In addition , eight 10µF 63VW bypass capacitors are connected across the battery input leads , following the 100A fuse. These capacitors effectively cancel out the indu ctance of the battery cables and th ereby provide full power to the transformer and Mosfets at the switching fre quency of 20kHz. The secon dary windings of the transformer con sist of four 9-turn windings wh ich are then connected to form two centre-tapped 18-turn windings in parallel. This arrange- ment minimises the leakage inductance of the transformer. Zener diodes ZD2-ZD5 provide protection for the Mosfets. ZD4 and ZD5 are 18V zeners which protect the gates of the Mosfets against over-drive. ZD2 and ZD3 are 75V zeners which protect the drains of the Mosfets from spikes as they switch off. These zeners clamp the drains to about 80V peak and thus prevent damage. The response time of this overvoltage clamp action is about lµs which is not really fast enough. To provide protection during this initial lµs period, we have specified Mosfets which have avalanche protection. This means that at the breakdown voltage of the Mosfet (100V), an internal zener provides protection for a short time until the external protection circuitry takes over. Voltage feedback As noted above, IC4 and optocou- pler IC5 are used to provide voltage feedback to the switchmode controller (ICl). IC4 and the optocoupler must be fully isolated from the 24V input supply, which means that they must be powered from the high voltage DC output. Thus, the 365V supply is fed to a network consisting of five series 10kQ dropping resistors plus a 15V zener diode, ZD6. This provides the 15V supply for IC4 and IC5. The voltage feedback network consists of a high-voltage 4. 7MQ resistor (Philips VR37) and a 12kQ resistor connected to 0V (ie, the 0V of the high voltage supply, not the 24V battery). IC4 monitors the voltage across the 12kQ resistor and drives the linear optocoupler, IC5. IC5 provides the necessary electrical isolation between input and output. This device has high linearity and this is due to the use of two internal photodiodes, one on the isolated side to supply the output (pins 5 & 6) and a second (pins 3 & 4) to provide feedback to the LED driver circuit. The isolated photodiode output at pins 5 & 6 of IC5 is connected between the+ 15V supply rail and pin 3 of IC3. The current from the photodiode develops a voltage across trimpot VRl which is amplified with a gain of about four by IC3. IC3 then feeds pin 16 of ICl and thus completes the feedback loop. Power supply The 24V input from the battery bank is connected via heavy duty cables (ie, starter motor cables) and a 100A/ 125A cartridge fuse to the centre tap of the transformer. Because of the high currents involved, there is no on/off switch for this main supply; the 24V . input is permanently applied to this part of the circuit. Switch Sl feeds 24V to the lowpower part of the circuitry and LED 1 indicates when the power is switched on. The 24V supply from switch Sl is fed via an 82Q 5W resistor and clamped against transient voltages using ZDl. T·he 24V supply then feeds the input ofa 15V regulator (REGl) which supplies the ICs. The 10µF capacitors at the input and output of the regulator are for supply decoupling. That's all for this month. In Pt. 3, we will describe the circuitry for sinewave generation and the H-pack switching output which converts the 365V DC to 240VAC. SC NovEMBEH 1992 31