Fullscreen HTML5Med Res (??MB)HTML4

Plastic Pow 175 watts into 4 ohms; 125 watts into 8 ohms This new amplifier module is a real powerhouse. It will deliver 175 watts into a 4Ω load or 125 watts into 8Ω loads for a rated distortion of .01%. It is very quiet, very stable and suitable for musical instruments or any hifi application. W E HAVE HAD THIS amplifier under development for a long time and now that the new Motorola MJL21193/94 series transistors have become available, we can finally publish it. These new bipolar power transistors can be considered to be the plastic replacements for the very popular MJ15003/4 TO-3 metal encapsu­lated transistors. As we see it, all TO-3 power transistors will eventually be phased out and so these new plastic transistors will become one of the standard power transistors in the future. And while plastic power transistors are usually not as rugged as their metal equivalents, these new 22  Silicon Chip Motorola MJL21193/94 transistors are exceptional in this regard. They are rated at 200 watts (<at> Tcase 25°C), 16 amps continuous collector cur­rent (30 amps peak) and 250 volts (Vceo). This compares with the MJ15003/4 series which are rated at 250 watts, 20 amps and 140 volts. This simple comparison might suggest that the latter devices are still more rugged but when you look at “second break­ down” characteristics, the ability of a transistor to handle high currents at high voltage, the new plastic transistors are clearly superior. Not only do they have a much higher collector voltage rat­ing, 200V versus 140V, they have Vcbo (collec- tor base voltage, open emitter) and Vcex (collector emitter voltage, base reverse biased) ratings of 400V and can deliver considerably more current than the TO-3 types when high voltage is applied. For example, with 100V between collector and emitter, the MJ15003/4 series can deliver 1A. By contrast, with the same voltage applied, the MJL21193/4 series can deliver about 1.7A, a considerable in­ crease. (Note: both these figures refer to a one-second non-repetitive pulse condition). As well, the new plastic power transistors feature higher current gain, a better current gain-bandwidth product (4MHz versus 2MHz) and wer! By LEO SIMPSON & BOB FLYNN lower distortion when used in class-B amplifier stages. All of these factors combine to enable an improved power amplifier design. In fact, when compared to our previous design featuring MJ15003/4 transistors – the Studio 200 published in the February 1988 issue – this new design delivers considerably more power. Fig.1 shows the load lines for 4Ω and 8Ω resistive loads in the new amplifier, together with reactive load lines for (2.83Ω + j2.83Ω) and (5.6Ω + j5.6Ω). Also shown on Fig.1 are concave maximum power hyperbolas showing the 400W rating for two Motorola MJL21193/4 transistors and the one-second SOAR curve. Actually, we have not shown Motorola’s full SOAR curve; it extends to 250V. As well as the performance advantage, the new plastic power transistors feature single hole mounting to a flat heatsink surface; there is no need for a heatsink flange or bracket as is the case with TO-3 power transistors. Performance Full details of performance are shown in the separate panel and the various power and frequency response plots. As noted above, the power rating is 175 watts into 4Ω and 125 watts into 8Ω at a rated total harmonic distortion of less than .01%. The music power outputs are 230 watts and 140 watts respectively, giving a headroom of 1.1dB for 4Ω loads and 0.4dB for 8Ω loads. However, this parameter is really a measure of the regulation of the power transformer and can be ignored. For a really good power supply, the music power and the continuous power ratings of any amplifier will be almost equal. As can be seen from the distortion curves of Figs.2, 3, 4 & 5, while we have quoted a rated distortion of .01%, the typical distortion of the amplifier is actually below .002%, depending on the frequency and power output. Also, for frequencies above 10kHz, and approaching full power, the distortion April 1996  23 rises above .01% to as high as .03%. The effects of this are inaudible though, since harmonics of 10kHz are above the range of human hearing. While we have rated the amplifier fairly conservatively, using .01% harmonic distortion as the benchmark for full power, if you drive the amplifier just to the point of clipping, say where the curve reaches 0.3% on Fig.5, the amplifier will deliver over 200 watts. This will naturally be boosted if the mains voltage is above 240VAC, as it normally is in urban areas. This amplifier module is also very quiet, as is expected from modern circuit design. The residual noise is better than -114dB unweighted (20Hz to 20kHz filter) or -122dB A-weighted. That is much quieter than any CD player! Fig.1: load lines for 4Ω and 8Ω resistive loads in the new ampli­fier, together with the arched reactive load lines for (2.83Ω + j2.83Ω) and (5.6Ω + j5.6Ω). The concave curves show the 400W power hyperbola (dotted) and the one-second SOAR curve, for two Motorola MJL21193/4 transistors. The module As can be seen from the photos, this amplifier module is assembled onto a reasonably compact PC board measuring 100 x 165mm, with the four output power transistors and three smaller power devices mounted along one edge for easy mounting to a vertical heatsink. The PC board has two supply fuses on board and provi­ sion for temporary mounting of two 5W wirewound resistors which are used for setting the quiescent current. We’ll have more to say about that later in the article. AUDIO PRECISION SCTHD-HZ THD+N(%) vs FREQ(Hz) 5 21 FEB 96 10:02:08 1 0.1 0.010 0.001 T T .0005 20 100 1k 10k 20k Fig.2: THD (total harmonic distortion plus noise) versus frequen­cy at 150W RMS into a 4Ω load. 24  Silicon Chip Circuit details The full circuit of the amplifier mod­ule is shown in Fig.7. For those who are familiar with previous power amplifier circuits we have published, this design is similar to the configuration of the 120W Mosfet amplifier we featured in November and December 1988. Superficially, all we have done is substitute bipolar output transistors for the Mosfets. In fact, there is a lot more to it than that as will become apparent as we describe the vari­ous circuit features. Which brings us to the point: why use bipolar transistors instead of Mos­ fets? The reasons are quite straightforward. While Mosfet output stages in amplifiers have the virtue of being rugged they are generally more expensive than equivalent bipolar power transistors. For a given circuit configuration and power supply, bipolars will always deliver more power. As well, they don’t need such large quiescent AUDIO PRECISION SCTHD-HZ THD+N(%) vs FREQ(Hz) 5 21 FEB 96 09:56:03 1 Model Railway Projects 0.1 0.010 0.001 T .0005 20 100 1k 10k 20k Fig.3: THD distortion versus frequency at 110W RMS into an 8Ω load. current in the output stage and that translates to less heat and again, more audio power output. Inevitably, some readers may question why we used the con­figuration of the November 1988 circuit rather than the well-proven Hitachi configuration featured in our December 1987 & February 1988 issues. In fact, we built up prototypes with both circuits. Both performed very well with the Hitachi circuit giving slightly less harmonic distortion at frequencies above 10kHz. However, the circuit featured in Fig.7 gave substantially more power before the onset of clipping and so it won out. Fifteen transistors and three diodes make up the semicon­ductor count of the circuit of Fig.7. The input signal is coupled by a 2.2µF capacitor and 1kΩ resistor to the base of Q1 which AUDIO PRECISION SCTHD-W THD+N(%) vs measured 10 LEVEL(W) 21 FEB 96 09:45:00 Available only from Silicon Chip Price: $7.95 (plus $3 for postage). Order by phoning (02) 979 5644 & quoting your credit card number; or fax the details to (02) 979 6503; or mail your order with cheque or credit card details to Silicon Chip Publications, PO Box 139, Collaroy, NSW 2097. ➦ Use this handy form 1 Enclosed is my cheque/money order for $________ or please debit my 0.1 ❏ Bankcard   ❏ Visa   ❏ Mastercard Card No: ______________________________ 0.010 Card Expiry Date ____/____ Signature ________________________ Name ___________________________ 0.001 Address__________________________ .0005 0.5 1 10 100 300 __________________ P/code_______ Fig.4: THD versus power at 1kHz into an 8Ω load. April 1996  25 AUDIO PRECISION SCTHD-W THD+N(%) vs measured 10 LEVEL(W) 21 FEB 96 09:47:03 1 0.1 0.010 0.001 .0005 0.5 1 10 100 300 Fig.5: THD versus power at 1kHz into a 4Ω load. AUDIO PRECISION SCFRQRES AMPL(dBr) vs FREQ(Hz) 5.0000 Vbe multiplier 21 FEB 96 09:51:55 4.0000 3.0000 2.0000 1.0000 0.0 -1.000 -2.000 -3.000 -4.000 -5.000 20 100 1k 10k 50k Fig.6: frequency response at 4W into an 8Ω load. to­gether with Q2 makes up a differential pair. Q3 is a constant current tail which sets the current through Q1 & Q2 and thereby makes the amplifier insensitive to variations in the power supply rails (this is known as PSRR; power supply rejection ratio). The collector loads of Q1 & Q2 are provided by current mirror tran­sistors Q4 & Q5. Commonly used 26  Silicon Chip of Q1 connects to the base of Q7, part of a cascode stage comprising Q7 & Q8, with Q6 pro­viding a constant current load to Q8. A 3.3V zener diode, ZD1, provides the reference bias to the base of Q8 (to see how a cascode circuit works, see the separate panel in this article). A 100pF capacitor from the collector of Q8 to the base of Q7 rolls off the open-loop gain of the amplifier to ensure a good margin of stability. The output signal from the cascode stage is coupled directly to the output stage, comprising driver transistors Q10 & Q11 and the four output transistors, Q12-Q13. Actually, it may look as though the collector of Q6 drives Q10 and that Q8 drives Q11, and indeed they do, but in reality, the signals to the bases of Q10 and Q11 are identical, apart from the DC offset provided by Q9. in operational amplifier ICs, current mirrors provide increased gain and improved linearity in differential amplifier stages. In a conventional direct-coupled amplifier, the signal from the collector of Q1 would be connected directly to the base of the following class-A driver stage transistor. In our circuit though, the signal from the collector Q9 is a “Vbe multiplier”. It can be thought of as a temper­ a turecompensated floating voltage source of about 2V. Q9 multi­plies the voltage between its base and emitter, as set by VR1, by the ratio of the total resistance between its collector and emitter (470Ω + 100Ω + VR1) to the resistance between its base and emitter (100Ω + VR1). In a typical setting, if VR1 is 100Ω (note: VR1 is wired as a variable resistor), the voltage between collector and emitter will be:     Vce = Vbe x 670/200 = (0.6 x 670)/200 = 2.01V In practice, VR1 is adjusted not to produce a particular voltage across Q9 but to set the quiescent current through the output stage transistors. We’ll describe setting the quiescent current later in this article. Because Q9 is mounted on the same heatsink as the driver and output transistors, its temperature is much the same as the output devices. This means that its base-emitter voltage drops as the temperature of the output devices rises and so it throttles back the quiescent current if the devices become very hot, and vice versa. Before leaving the cascode stage, we should mention the bias arrangements. As already noted, zener diode ZD1 sets the bias on the base of Q8, however the current through the cascode transistors is set by constant current source Q6 which has its base-emitter Fig.7: this direct coupled amplifier module uses a differential input stage (Q1,Q2) with a constant current tail (Q3) and current mirror load (Q4,Q5). This drives a cascode stage (Q7,Q8) with constant current load (Q6). Quiescent current in the output stage is set by VR1 and Q9. The output stage is a complementary class-AB Darlington configuration using Q10 and Q11 as the drivers and Q12 to Q15 as the power devices. bias set by the two diodes, D1 & D2. Because of D1 & D2, Q6 applies 0.62V to its emitter resistor and this thereby sets the current through Q6, Q8 & Q9 to 13mA. Note that D1 & D2 also provide the base-emitter bias to Q3 which sets the current through Q1 & Q2. Note too that although D1 & D2 provide identical bias to Q3 & Q6, Q3 applies a higher vol­ tage, 0.69V, to its 220Ω resistor. How can this be? The answer is partly that Q3 is operating at a slightly lower current (3mA rather than 13mA) but mainly because the BC556 transistors require less base-emitter voltage to turn them on than the BF470 used for Q6. Driver & output stages As already mentioned, Q10 & Q11 are the driver stages and they, like the output transistors, operate in classAB mode (ie, class B with a small quiescent current). Resistors of 100Ω are connected in series with the bases of these transistors as “stoppers” and they reduce any tendency of the output stages to oscillate supersonically. In order to deliver the high output currents required, four output transistors are used, essentially as paralleled pairs. Each pair, Q12/Q13 and Q14/ Q15, has its bases and collectors connected together and the emitters connected to the common­ed output via 0.47Ω 5W resistors. The resistors are includ­ed mainly to ensure a degree of current sharing between the transistors in each paralleled pair. For example, if the output stage was delivering 9 amps (possible at full power into a 4Ω load) and one transistor say, Q12, had twice the gain of Q13. The initial effect of this would be for Q12 to take twice as much current as Q13; ie, 6A versus 3A. However, if Q12 had 6A through it, its emitter resistor would have 2.82V across it and Q13’s emitter resistor would only have 1.41V across it. The net effect would be that the bias to Q12 would be throttled back substantially and so while Q12 would still take more current, the sharing would be April 1996  27 Cascode Operation Explained A cascode stage is one where two transistors are connected in series, as shown in Fig.8. This shows an idealised circuit with a precise reference voltage (Vref) applied to the base of Q2. In one sense, Q2 acts like an emitter follower and applies a fixed DC voltage (Vref - Vbe) to the collector of Q1. This con­stant supply voltage at the collector of Q1 eliminates any gain variations which would otherwise occur if Q1’s collector voltage was able to vary. The varying current drawn by Q1 due to its input signal then becomes the signal drive to the emitter of Q2. Because of the constant voltage at its base, Q2 is effectively connected much more even and so Q12 would not overheat. The emitter resistors also help to stabilise the quiescent current to a small degree and slightly improve the frequency response of the output stage by adding local current feedback. Negative feedback is applied from the output stage back to the base of Q2 via an 18kΩ resistor. The amount of feedback and therefore the gain, is set by the ratio of the 18kΩ resistor to the 820Ω value at the base of Q2. Thus the gain is 23. The low frequency rolloff is mainly set by the ratio of the 820Ω resistor to the impedance of the associated 100µF capacitor. This has a -3dB point of about 2Hz. The 2.2µF input capacitor and 18kΩ base bias resistor feed­ ing Q1 have a more important effect and have a -3dB point at about 4Hz. The two time-constants combined give an overall roll­off of -3dB at about 6Hz. Fig.8: an idealised cascode circuit. This has a precise reference voltage (Vref) applied to the base of Q2. At the high frequency end, the 820pF capacitor and the 1kΩ resistor feeding the base of Q1 form a low pass filter which rolls off frequencies above 195kHz (-3dB). The overall amplifier fre­quency response can be seen in the diagram of Fig.6. An output RLC filter comprising a 6.8µH choke, a 6.8Ω resistor and a 0.15µF capacitor couples the output signal of the amplifier to the loudspeaker. It isolates the amplifier from any large capacitive reactances in the load and thus ensures stabili­ty. It also helps attenuate RF signals picked up by the loud­speaker leads and stops them being fed back to the early stages of the amplifier where they could cause RF breakthrough. The low pass filter at the input is also there to prevent RF signal breakthrough. Finally, before leaving the circuit description, we should note that the PC board itself is an integral part of Fig.9: suggested power supply for the amplifier. This should be upgraded if the amplifier is to be used with 4Ω loads, with 20,000µF (2 x 10,000µF) on each supply rail. 28  Silicon Chip as a “grounded base” stage and it converts the varying signal current at its emitter to a signal voltage at its collector. The combined effect of operating Q1 with a constant collec­tor voltage and Q2 in grounded base mode gives a stage with much improved linearity and bandwidth compared with a single common emitter stage. Cascode stages are a common feature of RF circuitry where their wide bandwidth is desirable. Cas­ code stages were originally designed around valves and the word “cas­code” is derived from the phrase “cas­­cad­ed via the cathode”, a reference to the cathode of a valve. the circuit and is a major factor in the overall performance. The board features star earthing, for minimum interaction between signal, supply and output currents. Note that the small signal components are clustered at the front of the board while all the heavy current stuff is mostly at the back and sides. For good tempera­ ture compensation of the quiescent current, all the output tran­sistors, the driver transistors and the Vbe multiplier, Q9, are mounted on the same heatsink. Suggested power supply Fig.9 shows the circuit of a suggested power supply for the amplifier. Note that we regard this as a “minimum spec” power supply and one which should be upgraded if the amplifier is to be used with 4Ω loads. If this is the case, we suggest that 20,000µF (2 x 10,000µF) on each supply rail would be the minimum required, in order to satisfy the ripple current demands when the amplifier is delivering high power. The power transformer is a 300VA toroidal type which may seem rather large but remember that this amplifier will easily deliver more than 200 watts at the onset of clipping and there­fore needs a 300VA transformer, particularly if it is to be used in professional sound reinforcement applications. The power supply and the amplifier module will need to be mounted in Fig.10: install the components as shown here, taking care to ensure that all polarised parts are correctly oriented. Note that the 5W resistors are mounted slightly proud of the board. a substantial metal case with a large heatsink. The bridge rectifier will need to be mounted on the metal chassis because it will dissipate quite a large amount of heat when the amplifier is delivering high power. the amplifier is in­tended for continuous full power delivery at frequencies above 10kHz, then the 6.8Ω resistor in the output filter should be a wire­ wound type with a rating of at least 5W, otherwise it will burn out. Choke L1 is wound with 24.5 turns of 0.8mm enamelled copper wire on a 13mm plastic former. Alternatively, some kitset suppliers will provide this choke as a finished component. When installing the fuse clips, note that they each have little lugs on one end which stop the fuse from moving. If you install the clips the wrong way, you will not be able to fit the fuses. Board assembly The component overlay diagram of the PC board is shown in Fig.10. Before starting board assembly, it is wise to check the board carefully for open or shorted tracks or undrilled lead holes. Fix any defects before fitting the components. Start by inserting the PC pins and the resistors. When in­stalling the diodes, make sure that they are inserted with the cor­rect polarity and that you don’t confuse D1 & D2 (1N914 or 1N4148) with the 3.3V zener diode (BZX79-C3V3 or equivalent). Take care when installing the electrolytic capacitors to make sure that they are installed the right way around. Note that the 100pF compensation capacitor from the collec­tor of Q8 to the base of Q7 should have a voltage rating of at least 100V while the 0.15µF capacitor in the output filter should have a rating of 400V. Another point to be noted is that if Both Q6 and Q8, which are BF470 and BF469 respectively, are fitted with U-shaped flag heatsinks, as shown here. April 1996  29 Fig.11: this diagram shows the heatsink mounting details for the power transistors. After mounting, use an ohmmeter to confirm that each device has been correctly isolated from the heatsink (there should be an open circuit between the heatsink and the device collectors). The 560Ω 5W wirewound resistors can also be installed at this stage; they are wired to PC stakes next to each fuseholder and are used during the setting of quiescent current. Next, mount the smaller transistors; ie, BC546, BC556, BF469 and BF470. Both Q6 & Q8 need to be fitted with U-shaped heat­ sinks, as shown in Fig.10. The four output transistors, the driver transistors (Q10 & Q11) and the Vbe multiplier Q9 are mounted vertically on one side of the board and are secured to the heatsink with 3mm machine screws. Perhaps the best way of lining up the transistors before they are soldered to the board is to temporarily attach them to the heatsink (don’t bother with heatsink compound or washers at this stage). This done, poke all the transistor leads through their corresponding holes in the board and line up the board so that its bottom edge is 10mm above the bottom edge of the heatsink. This ensures that the board will be horizontal when fitted with 10mm spacers at its front corners. Note that you will have to bend out all the transistor leads by about 30°, in order to poke them through the PC board. The heatsink will need to be drilled and tapped to suit 3mm machine screws. The relevant drilling details are shown in Fig.12. You can now solder all the transistor leads to the PC board. Having done that, undo the screws attaching the transis­tors to the heatsink and then fit mica washers and apply heatsink compound to the transistor mounting surfaces and the heatsink areas covered by the mica washers. The details for mounting these transistors are shown in Fig.11 . Alternatively, you can dispense with mica washers and heatsink compound and use silicone impregnated thermal washers instead, as can be seen in the pho­tos. Whichever method you use, do not over-tighten the mount­­ing screws. PARTS LIST 1 PC board, code 01104961, 100mm x 165mm 4 20mm fuse clips 2 20mm 5A fuses 1 coil former, 24mm OD x 13.7mm ID x 12.8mm long, Philips 4322 021 30362 2 metres 0.8mm diameter enamelled copper wire 7 PC board pins 1 large single sided heatsink, Jaycar Cat. HH-8546 or equivalent 2 TO-126 heatsinks, Altronics Cat. H-0504 or equivalent 4 TO-3P insulating washers (for output transistors – see text) 3 TO-126 insulating washers 4 3mm x 20mm screws 3 3mm x 15mm screws 7 3mm nuts 1 200Ω trimpot Bourns 3296W series (VR1) 30  Silicon Chip Semiconductors 2 MJL21194 NPN power transistors (Q12,Q13) 2 MJL21193 PNP power transistors (Q14,Q15) 2 MJE340 NPN driver transistors (Q9,Q10) 1 MJE350 PNP driver transistor (Q11) 1 BF469 NPN transistor (Q8) 1 BF470 PNP transistor (Q6) 3 BC546 NPN transistors (Q4, Q5,Q7) 3 BC556 PNP transistors (Q1, Q2,Q3) 2 1N914 diodes (D1,D2) 1 3.3V 0.5W zener diode (ZD1) Capacitors 4 100µF 63VW electrolytic 1 100µF 16VW electrolytic 1 2.2µF 16VW electrolytic 1 0.15µF 400V MKC, Philips 2222 344 51154 or Wima MKC 4 5 0.1µF 63V MKT 1 820pF 50V ceramic 1 100pF 100V ceramic Resistors 4 0.47Ω 5W 2 560Ω 5W (for current setting) 1 15kΩ 1W 1 5.6kΩ 1W 1 6.8Ω 1W 2 18kΩ 0.25W 1 6.8kΩ 0.25W 1 1kΩ 0.25W 1 820Ω 0.25W 1 470Ω 0.25W 3 220Ω 0.25W 1 180Ω 0.25W 2 150Ω 0.25W 3 100Ω 0.25W 1 68Ω 0.25W 1 47Ω 0.25W Fig.12: this diagram shows the drilling details for the large finned heatsink. April 1996  31 Fig.13: this is the full-size etching pattern for the PC board. Check the board carefully for defects before installing any parts. Now check with your multimeter, switched to a high Ohms range, that there are no shorts between the heatsink and any of the transistor collector leads. If you do find a short, undo each transistor mounting screw until the short disappears. It is then a matter of locating the cause of the short and re­mounting the offending transistor. Double-check all your soldering and assembly work against the circuit of Fig.7 and the component layout diagram of Fig.10. Set trimpot VR1 fully anticlockwise so that it is at minimum resistance. Remove both fuses and ensure that the 560Ω 5W resis­ tors are wired across both fuseholders, as described above. Testing We will assume that you have made or have access to a suit­ able power supply which is already working. That being the case, connect the supply rails and apply power. No loudspeaker or resistive load should be connected at this stage. Check the voltages shown on the circuit of Fig.7. These measurements were made with an AC supply voltage of 240VAC. If your mains voltage is PERFORMANCE Output power....................... 125 watts into 8Ω; 175 watts into 4Ω Music power........................ 140 watts into 8Ω; 230 watts into 4Ω Frequency response............ -0.3dB down at 20Hz and 20kHz (see Fig.6) Input sensitivity.................... 1.37V RMS (for full power into 8Ω) Harmonic distortion............. <.03% from 20Hz to 20kHz; typically <.003% Signal-to-noise ratio ����������� 114dB unweighted (20Hz - 20kHz); 122dB A-weighted Damping factor.................... >95 at 100Hz & 1kHz; >50 at 10kHz. Stability................................ Unconditional 32  Silicon Chip higher, and this will normally be the case, then the amplifier supply rails will be increased accordingly. Now measure the voltage at the output of the amplifier. It should be within ±50mV of 0V. If it is not close to zero, switch off the power as you have a fault. Check the voltages in the early stages as this should give you a guide to where the fault lies. The things to look for include: missed solder connections; solder splashes between tracks; incorrectly connected transistors; incorrect transistor types; and parts in the wrong way around, etc. Now monitor the voltage across one of the 560Ω 5W resis­tors. With VR1 fully anticlockwise, the voltage should be close to zero since there is no quiescent current in the output stage. Now slowly wind VR1 clockwise until the voltage starts to rise. Set VR1 for a voltage of 14V across the 560Ω resistor. This is equivalent to a quiescent current of 25mA or 12.5mA through each output transistor. You can check this by measuring the voltage drop across any of the 0.47Ω 5W emitter resistors. The average value across all four resistors should be 11mV. Leave the amplifier to run for 10 minutes or so and then retouch the setting of VR1 if necessary. Finally, fit the 5A fuses and the SC module is finished.