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SILICON
CHIP
If you are seeing a blank page here, it is
more than likely that it contained advertising
which is now out of date and the advertiser
has requested that the page be removed to
prevent misunderstandings.
Please feel free to visit the advertiser’s website:
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Vol.9, No.8; August 1996
Contents
FEATURES
4 Electronics On The Internet
You can check out data sheets on the latest devices, order parts or even market
products via the World Wide Web – by Sammy Isreb
64 Cathode Ray Oscilloscopes, Pt.4
It’s easy to get the wrong results when using an oscilloscope. Here’s how to
ensure that your measurements are always accurate – by Bryan Maher
76 An Introduction To IGBTs
ELECTRONIC STARTER FOR
FLUORESCENT LIGHTS – PAGE 14
Insulated gate bipolar transistors (IGBTs) combine the best characteristics of
bipolar transistors and Mosfets in one package. Here’s a look at how these
devices work – Motorola Semiconductor
PROJECTS TO BUILD
14 Electronic Starter For Fluorescent Lights
It’s built into a standard starter case and provides rapid starting. Fit this to your
fluorescent lights and get rid of the blinkety-blink-blink-blinks – by John Clarke
20 Build A VGA Digital Oscilloscope; Pt.2
Second article has all the circuit details – by John Clarke
30 A 350-Watt Audio Amplifier Module
Uses eight plastic Mosfets, is easy to build and delivers 350W into 4-ohms or
200W into 8-ohms – by Leo Simpson
54 Portable Masthead Amplifier For TV & FM
Are your TV signals weak and noisy? This masthead amplifier can mean the
difference between a lousy picture and a good picture – by Branco Justic
340 WATT AUDIO AMPLIFIER
MODULE – PAGE 30
SPECIAL COLUMNS
38 Satellite Watch
What’s available on satellite TV – by Garry Cratt
40 Serviceman’s Log
How many symptoms from one fault? – by the TV Serviceman
72 Radio Control
Multi-channel radio control transmitter; Pt.7 – by Bob Young
82 Computer Bits
Customising the Win95 desktop & start menus – by Greg Swain
86 Vintage Radio
A rummage through my junk – by John Hill
DEPARTMENTS
2 Publisher’s Letter
8 Circuit Notebook
19 Order Form
90 Product Showcase
93 Ask Silicon Chip
95 Market Centre
96 Advertising Index
MASTHEAD AMPLIFIER/
ACTIVE ANTENNA – PAGE 54
August 1996 1
Publisher & Editor-in-Chief
Leo Simpson, B.Bus., FAICD
Editor
Greg Swain, B.Sc.(Hons.)
Technical Staff
John Clarke, B.E.(Elec.)
Robert Flynn
Rick Walters
Reader Services
Ann Jenkinson
Advertising Manager
Christopher Wilson
Phone (02) 9979 5644
Mobile 0419 23 9375
Regular Contributors
Brendan Akhurst
Garry Cratt, VK2YBX
Julian Edgar, Dip.T.(Sec.), B.Ed
John Hill
Mike Sheriff, B.Sc, VK2YFK
Philip Watson, MIREE, VK2ZPW
Bob Young
Photography
Stuart Bryce
SILICON CHIP is published 12 times
a year by Silicon Chip Publications
Pty Ltd. A.C.N. 003 205 490. All
material copyright ©. No part of
this publication may be reproduced
without the written consent of the
publisher.
Printing: Macquarie Print, Dubbo,
NSW.
Distribution: Network Distribution
Company.
Subscription rates: $54 per year
in Australia. For overseas rates, see
the subscription page in this issue.
Editorial & advertising offices:
Unit 34, 1-3 Jubilee Avenue, Warrie
wood, NSW 2102. Postal address:
PO Box 139, Collaroy Beach, NSW
2097. Phone (02) 9979 5644. Fax
(02) 9979 6503.
PUBLISHER'S LETTER
New technology
marches on
This month we have a number of articles
which help signpost the relentless march of
technology. The first of these is the article
entitled “Electronics on the Internet” starting on page 4. This gives a glimpse of the
huge amount of information pertaining to
electronics which is now available on the
Internet. Some people are very wary of the
hyperbole surrounding the “net” but there is
no denying that large numbers of companies, organisations and individuals
are becoming involved in it.
Somewhat more prosaic perhaps, is the article on the Electronic Starter
for fluorescent lamps, beginning on page 14. The notable point about this
circuit is not so much that the integrated circuit at its heart is a clever chip
but that it is only available in surface-mount form. Increasingly, many new
ICs are only available in this format, so if you don’t already have a set of
prescription close-up spectacles, it is only a matter of time before you will
need them.
Interestingly, if the starter IC had been a conventional 8-pin DIL package it
would have been too big. You can expect to see this circuit as a commercial
product within a year or so.
On page 30, we are featuring a new high power amplifier module using
plastic power Mosfets from England. The point of interest is not that they
come from England but that we have probably now seen the last of power
transistors or Mosfets in metal TO-3 cans; plastic rules supreme.
On page 64, we have the second article on our VGA Oscilloscope and
this highlights the fact that VGA monitors for personal computers, the latest
and the greatest in display technology only a few years ago, are now being
discarded in large numbers as people upgrade their computers.
While the oscilloscope project is a good application for these otherwise
unused and unloved computer monitors, it seems to us that many people are
rushing headlong into the purchase of new computers without ever having
fully come to grips with the capabilities of their older machines.
Finally, on page 76, we feature an article on IGBTs. These hybrid devices,
a marriage between Mosfets and bipolar transistors, first appeared about 10
years ago but have been largely confined to heavy power tasks such as traction
motor control. They are gradually making their way into more general use
and indeed they were featured in the SILICON CHIP 2kW sinewave inverter
in 1992. In the future, you can expect to see them in car ignition systems,
in audio amplifiers and in most general power applications.
Leo Simpson
ISSN 1030-2662
WARNING!
SILICON CHIP magazine regularly describes projects which employ a mains power supply or produce high voltage. All such projects should
be considered dangerous or even lethal if not used safely. Readers are warned that high voltage wiring should be carried out according to the
instructions in the articles. When working on these projects use extreme care to ensure that you do not accidentally come into contact with
mains AC voltages or high voltage DC. If you are not confident about working with projects employing mains voltages or other high voltages,
you are advised not to attempt work on them. Silicon Chip Publications Pty Ltd disclaims any liability for damages should anyone be killed
or injured while working on a project or circuit described in any issue of SILICON CHIP magazine. Devices or circuits described in SILICON
CHIP may be covered by patents. SILICON CHIP disclaims any liability for the infringement of such patents by the manufacturing or selling of
any such equipment. SILICON CHIP also disclaims any liability for projects which are used in such a way as to infringe relevant government
regulations and by-laws.
Advertisers are warned that they are responsible for the content of all advertisements and that they must conform to the Trade Practices Act
1974 or as subsequently amended and to any governmental regulations which are applicable.
2 Silicon Chip
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unit suitable for use with directional parabolic reflectors
etc. PCB: 63 x 37mm: $10 (K64).
FLUORESCENT LIGHT HIGH FREQUENCY BALLASTS
European made, new, “slim line” cased high frequency
(HF) electronic ballasts. They feature flicker free starting,
extended tube life, improved efficiency, no visual flicker
during operation (as high frequency operation), reduced
chance of strobing with rotating machinery, generate no
audible noise and generate much reduced radio frequency
interference compared to conventional ballasts. Some
models include a dimming option which requires either
an external 100kΩ potentiometer or a 0-10V DC source.
Some models require the use of a separate filter choke
(with dimensions of 16 x 4 x 3.2cm) - this is supplied
where required. We have a limited stock of these and are
offering them at fraction of the cost of the parts used in
them! Type B: 1 x 16W tube, dimmable, filter used, 43 x
4 x 3cm: $16. Type F: 1 x 32W or 36W tube, dimmable,
no filter, 34 x 4 x 3cm: $18
(Cat G09, specify type).
27MHz RECEIVER CLEARANCE
Soiled 27MHz telemetry receivers. Enclosed in waterproof
die cast metal boxes, telescopic antenna supplied. 270 x 145
x 65mm. 2.8kg. Two separate PCBs. Receiver PCB has audio
output. Signal filter/squelch PCB is used to detect various
tones. Circuit provided: $12.
40-CHANNEL FM MICROPHONE
A hand held crystal locked 40-channel FM transmitter
with LCD display: 88-92MHz in 100kHz steps, 50m
transmission range. Perfect for use with synthesized FM
receivers: $50.
SPEED CONTROLLED GEARED MOTOR
Experiment with powering small vehicles, large children’s
cars, garage door openers, electric wheelchairs, rotisseries,
etc. etc. We supply a speed control PCB and components
kit, A 25A MOSFET and a 30A diode (flyback), and a used
12V geared windscreen wiper motor for a total price
of: $30.
CHARACTER DISPLAYS
We are offering three types of liquid crystal character
displays at bargain prices. The 40 x 2 character display
(SED1300F) is similar to the Hitachi 44780 type but is not
directly compatible. We will also have similar displays - data
available for a 16 x 4 and 32 x 4 display. Any mixture of
these displays is available for a crazy price of $22 each
or 4 for $70.
IR TESTER USING IR CONVERTER TUBE
Convert infra red into visible light with this kit. Useful
for testing infra red remote controls and infra red laser
diodes. We supply a badly blemished IR converter tube
with either 25 or 40mm diameter fibre optically coupled
input and output windows and our night vision high
voltage power supply kit, which can be powered from a
9V battery. These tubes respond to IR and visible light. A
very cheap IR scope could be made with the addition of
a suitable casing and objective lens and eyepiece. $30.
MISCELLANEOUS ITEMS
2708 EEPROMS: $1 each; 4164 MEMORY ICs: 16 for $10:
AC MOTOR, 1RPM Geared 24V-5W Synchronous motor plus
a 0.1 to 1RPM driver kit to vary speed, works from 12V DC:
$12 K38 + M30; SPRING REVERB, 30cm long with three
springs: $30 A10; MICROSONIC MICRO RECORD PLAYER,
Includes amplifier: $4 A11; LARGE METER MOVEMENTS:
moving iron, 150 x 150mm square face, with mounting
hardware: $10.
REFLECTIVE TAPE
High quality Mitsubishi brand all weather 50mm wide red
VHF MODULATOR KIT
For channels 7 and 11 in the VHF TV band. This is designed
for use in conjunction with monochrome CCD cameras to
give adequate results with a cheap TV. The incoming video
simply directly modulates the VHF oscillator. This allows
operation with a TV without the necessity of connecting
up wires, if not desired, by simply placing the modulator
within about 50cm from the TV antenna. Suits PAL and
NTSC systems. PCB: 63 x 37mm: $12 (K63).
‘MIRACLE’ ACTIVE AM ANTENNA KIT
Available soon. To be published in EA. After the popularity
of our Miracle UHF/VHF antenna kits we have produced
this AM version for our ‘Miracle’ series. Large antennas
are not the most attractive inside a house but sometimes
this is needed to receive a weak radio signal. This kit
will connect to a remote loop of wire, preferably outside
where reception is good, via coax cable and allow it to be
tuned from inside via varactor diodes. Radio reception is
greatly improved and it can even pickup remote stations
that a radio can’t receive with its ferrite rod antenna. No
connections are required to the existing radio as the
receiving end is coupled to the ferrite rod in the radio
with a loop of wire around the radio. Excellent kit for
remote country areas where radio reception isn’t very
good, or where a large antenna is not possible. Great for
caravanners, boats that venture far out to sea, etc. 2 x
PCBs and all on-board components.
BATTERY CHARGER WITH MECHANICAL TIMER
Simple kit which is based on a commercial 12 hour mechanical timer switch which sets the battery charging period from
0 to 12 hrs. Employs a power transistor and five additional
components. Can easily be “hard wired”. Information that
shows how to select the charging current is included. We
supply information, circuit and wiring diagram, a hobby box
with aluminium cover that doubles up as a heatsink, a timer
switch with knob, a power transistor and a few other small
components to give you a wide selection of charge current.
You will also need a DC supply with an output voltage which
is greater by about 2V than the highest battery voltage you
need to charge. As an example a cheap standard car battery
charger could be used as the power source to charge any
chargeable battery with a voltage range of 0-15V. Or you
could use it in your car. No current is drawn at the end of
the charging period: $15.
AUTOMATIC LASER LIGHT SHOW KIT
Kit as published in Silicon Chip May 96 issue. The display
changes every 5 - 60 seconds, and the time is manually
adjustable. For each of the new displays there are 8 different
possible speeds for each of the 3 motors, one of the motors
can be reversed in rotation direction, and one of the motors
can be stopped. There are countless possible interesting
displays which vary from single to multiple flowers, collapsing circles, rotating single and multiple ellipses, stars, etc.
etc. Kit makes an excellent addition to any lightshow and all
these patterns are enhanced by the use of a fog machine.
Kit includes PCB, all on board components, three small
DC motors, 3 high quality/low loss dichroic mirrors: $90.
Suitable 12V DC plugpack: $14.
LASER LIGHTSHOW PACKAGE
Our 12V Universal inverter kit plus a used 5mW+ helium-neon laser tube head plus a used Wang power supply
plus an automatic laser light show kit with dichroic mirrors
(as above): $200.
ARGON-ION HEADS
Used Argon - Ion heads with 30-100mW output in the blue
- green spectrum. Head only supplied. Needs 3Vac <at> 15A
for the filament and approx 100Vdc <at> 10A into the driver
circuitry that is built into the head. We provide a circuit for a
suitable power supply the main cost of which is for the large
transformer required: $170 from the mentioned supplier.
Basic information on power supply provided. Dimensions:
35 x 16 x 16cm. Weight: 5.9kg. 1 year guarantee on head.
Price graded according to hours on the hour meter: We have
had no serious problems with any of these heads as they
were used at a very low current in their original application.
Argon heads only: $300.
SIREN USING SPEAKER
Uses the same siren driver circuit as in the “Protect anything alarm kit”. 4-inch cone / 8-ohm speaker is included.
Generates a very loud and irritating sound with penetrating
high and low frequency components. Output has frequency
components between 500Hz and 4kHz. Current consumption
is about 0.5A at 12V. PCB: 46 x 40mm. As a bonus, we
include all the extra PCBs as used in the “Protect anything
alarm kit”: $12.
DC MOTORS
We have good stocks of the following high quality DC motors.
These should suit many industrial, hobby, robotics and
other applications. Types: Type M9 : 12V. I no load = 0.52A
<at> 15800 RPM at 12V. Weight: 150g. Main body is 36mm
diameter. 67mm long: $7 (Cat M9) Type M14 : Made for slot
cars. 4 to 8V. I no load = 0.84A at 6V. At max. efficiency I
= 5.7A <at> 7500 RPM. Weight: 220g. Main body diameter is
30mm. 57mm long: $7 (Cat M14).
ULTRASONIC COMMUNICATOR KIT
Ref: EA Sep/Oct 93. Signals picked up by an electret
microphone are modulated onto an oscillator which
drives a 40kHz ultrasonic transducer. This is received by
a 40kHz ultrasonic receiving transducer and is amplified
and detected. The detected signal is amplified by a simple
three transistor amplifier to drive a speaker. This makes a
communications link using ultrasound which can transmit
over a few metres. The quality of the sound is limited by
the narrow bandwidth of the transducers but this is an
interesting experiment. Both transmitter and receiver PCBs
are 63 x 33mm: $16 (K45).
BOG DEPTH SOUNDER KIT
Detect the presence and depth of any body filler on your
car. This simple circuit uses an oscillator which is oscillating
weakly. When steel is placed near the small search coil the
inductance shifts and the oscillator components are arranged
so the oscillator will stop running. The remainder of the
circuit simply detects when the oscillator stops and gives a
visual or audible indication of this. The circuit is arranged so
that the change in inductance needed to stop the oscillator
can be varied. This allows variable depth of filler sensing,
between 0 and about 3mm. Large areas of body filler over
3mm thick are generally considered undesirable as the filler
may lift or crack. Kit supplied includes pre-wound search
coil (33 x 22 x 10mm). A LED is supplied in the kit as the
visual indication. An audible indication can be obtained by
using a low power piezo buzzer, which is recommended but
not supplied with the kit: $12 (K62).
$2 for optional low power piezo buzzer.
HIGH VOLTAGE AC DRIVER
This kit produces a high frequency high voltage AC output
that is suitable for ionizing most gas filled tubes up to 1.2m
long. It will partially light standard fluorescent tubes up
to 1.2m long with just 2 connections being made, and
produce useful white light output whilst drawing less than
200mA from a 12V battery. Great for experimenting with
energy efficient lighting and high voltage gas ionization.
PCB plus all on board components, including high voltage
transformer: $18.
PC CONTROLLED PROGRAMMABLE
POWER SWITCH MODULE
This module is a four-channel programmable on/off timer
switch for high power relays. The timer software application
is included with the module. Using this software the operator
can program the on/off status of four independent devices
in a period of a week within a resolution of 10 minutes.
The module can be controlled through the Centronics or
RS232 port. The computer is opto-isolated from the unit.
Although the high power relays included are designed for
240V operation, they have not been approved by the electrical
authorities for attachment to the mains. Main module: 146
x 53 x 40mm. Display panel: 146 x 15mm. We supply: two
fully assembled and tested PCBs (main plus control panel),
four relays (each with 3 x 10A / 240V AC relay contacts),
and software on 3.5-inch disk. We do not supply a casing
or front panels: $92 (Cat G20).
August 1996 3
ELECTRONICS
ON THE
There’s a wealth of information on
the Internet for electronics designers
and engineers. You can check data
sheets on the latest devices, order
parts or even market your products
via the World Wide Web.
By SAMMY ISREB
Before you turn the page thinking
that this is just another Internet article
WAIT! It has no resemblance to the
myriad of articles on how to get onto
the Internet and use it. In fact, before
you can use any of the information
in this article you have to be already
“hooked up” to the Internet and be
familiar with its use.
So now that we know what this article is not about it is time to explain
what it is about. The Internet contains
hundreds of web sites and news groups
that are dedicated to electronics. The
news groups are made up chiefly of
support or help groups, while the
web sites are usually commercially
orientated.
Web sites
Fig.1: the National Semiconductor homepage screen. Among other things, it
allows the user to search for a component by part number, to browse a library
of datasheets and to seek out technical advice and sales information.
4 Silicon Chip
Many electronic companies have
embraced the World Wide Web with
open arms, setting up their own home
pages. The most valuable web sites
for designing circuits are those of the
IC manufac
turers. Many companies
include datasheets on all ICs that they
manufacture, as well as application
circuits and pricing and availability
data. Some web sites, such as National Semiconductor’s, even allow the
user to search for a device name and/
or number to find a datasheet for an
unknown device.
This service is very handy when
you have an unknown IC and want
to get more information on it. All datasheets and application notes are in
the *.pdf format and require a reader
such as Adobe Acrobat Reader Ver.
2.1 or later.
This software can be downloaded
from the National Semiconductor or
Adobe web sites. It will be compressed
in the pkzip format and is easily ex-
panded using any pkunzip compatible
archiving utility.
A feature of some IC manufacturer
web sites, such as the Motorola Semiconductor site, is a fax back feature.
With this, the user can request that
a datasheet be faxed to his/her fax
machine by simply selecting the
datasheet required and entering the
phone number. Failing this, some
web sites offer a “snail mail” service
which allows the selected information
to be mailed to the address entered at
the prompt.
With the microcontroller industry
booming, the Internet can be the best
way to select a family of chips to use.
Most manufacturers are represented
on the Internet. This allows easy
comparison with opposition systems
so that the one that best suits your
needs can be selected. After browsing
the Parallax web site, I decided that
their Basic Stamp looked like a fun
toy and ordered one. Be warned – lock
away your wallet before browsing web
pages!
As well as the microcontroller sites
that are run by the manufacturers,
there are quite a few sites that are conducted by electronics enthusiasts who
have fallen “in love” with a particular
type of microcontroller. These sites
usually have a deluge of circuits using
the microcontroller in question, with
some of the circuits being quite novel
and ingenious.
Probably the best feature of these
sites, however, is that their owners
are usually happy to answer any
questions/problems relating to their
particular microcontroller. Links to the
best of these sites are sometimes also
included in the manufacturer’s sites.
Fig.2: the National Semiconductor IC data page menu. You can download
selected information and even download the Adobe Acrobat reader if you
don’t already have a copy.
Circuit databases
There are also many web sites and
ftp (file transfer proto
col) servers
which contain databases of small circuits, along with short descriptions.
Many of these circuits are quite novel,
while some are downright strange in
their uses of standard components.
The sites also contain the standard
boring building block cir
cuits such
as two trillion uses for a 555 timer
and so forth.
Along with such sites, which are
usually operated by uni
versities or
electronics enthusiasts, are similar
sites run by commercial companies, or
for commercial companies by workers
or private enthusiasts. The Parallax ftp
Fig.3: this National Semiconductor data page shows information on the
LMX2216 low-noise amplifier/mixer IC. Clicking on the download icon allows
the datasheet to be downloaded.
site contains quite a few circuits using
their microcontrollers, as well as links
to similar sites.
Shareware
Electronic related shareware is
available freely on the Internet. Indeed, many ftp and web sites dedicat-
ed to electronics will have several directories containing useful programs.
These are usually split into various
categories such as PC board design and
manufacture, schematic plotting and
CAD programs, simulation programs
of various types, and educational files.
All of these files will be archived in
August 1996 5
Fig.4: the Parallax homepage. It offers information on the Basic Stamp and PIC
microcontrollers, includes quite a few circuits, and has links to various other
sites that offer circuits.
some way and can be downloaded in
the standard fashion.
As well as these shareware programs, many commercial soft
ware
manufacturers place demonstration
packages of their software on web/ftp
sites. This type of software is usually
either a “crippled” version of the real
thing or a much older superseded
version.
In either case, this software is good
enough for demonstration purposes
and gives the user the opportunity to
“try before buying”.
Mail order stores
As the Internet’s popularity increases, many mail order electronics
stores are setting up web sites on
which users can browse their goods
catalog and even select, order and pay
for their goods. While this trend is a
bit sluggish taking off in Australia,
there are many electronics dealers
overseas who sell goods via the World
Wide Web.
The fact that a price for the same
component can be obtained from dozens of shops from around the world
in a few minutes really allows for
customers to come out on top.
A tip to buying goods is to know
what you are looking for from the outset and do a web search on the item.
If you are uncomfortable giving credit
card details to an unknown company
halfway around the world, send them
a cheque by “snail mail” instead.
It is also now possible to consult
with several PC board design and
production companies on the web.
Whilst these business transactions
can’t be carried out fully online, it is
possible to get some idea of what the
various companies have to offer, what
type of pricing to expect, and previous
examples of their work.
Marketing
Once your perfect circuit is up and
running, the Internet is a great way to
get it onto the market. It is possible to
obtain quotes on various components
in high quantities from specialist
wholesale dealers and you can even
try negotiating a deal to supply the
finished product to one of the web’s
electronics dealers. An alternative to
this is to set up your own web site to
advertise your products.
If you do decide that your own web
page is the way to go, a good way to
increase the number of hits it receives
is to convince an established electronics web site to include your site as one
of its links.
News groups
Fig.5: the Motorola Semiconductor Products homepage. A feature of the
Motorola site is a fax-back facilitity.
6 Silicon Chip
Those who encounter a brick wall
when designing or repair
ing a circuit should try the electronics news
groups. A search of “electronics” will
give a list of several different news
groups dedicated to divergent disciplines of electronics. Make sure that
the one you select is the relevant one
for your problem, as not doing so can
make the users of the newsgroup you
mail quite irate.
After selecting the appropriate
news group, post a letter stating your
problem clearly. Be sure to include
your email address so that the solution
Web sites To Try
The web sites listed below are
among my favourites and are invaluable for any electronics work.
Note, however, that they represent
just a tiny fraction of the web sites
that deal with electronics – there
are thousands of others.
http://www.hitechsurplus.com/
High-tech surplus goods for sale
from North America
http://www.natsemi.com/
National Semiconductor homepage
http://www.parallaxinc.com/
Parallax homepage
Fig.6: this screen shows the Motorola datasheet page. You can ask questions by
clicking the Tech Support icon and then filling in the on-screen form.
to your problem can be forwarded to
you. After posting your problem, read
through a few other people’s problems
and try to solve them as this keeps the
news groups going.
Although the advice given on news
groups is very helpful, with many
electronic professionals giving advice, the advice given should never
be taken as gospel. Also, while most
conversations that go on in the news
groups deal with standard electronic
problems and can be very educational,
readers should be very wary of some
of the topics discussed.
On the day I accessed the news
groups to research this article, 10 of
the postings dealt with a person who
wanted to construct a 1,000,000V tesla
coil in his garage after seeing a tesla
coil on a television science show. Most
of the advice told him not to attempt it
if he valued his life, as it was not the
type of experiment to be attempted by
a beginner.
Another 18 postings dealt with a
student who wanted information on
constructing an electromagnetic gun
to fire metal stakes!
quite well supported, with web pages
containing information on the elec
tronics involved. There is also a model
rocketry newsgroup where problems
are quickly solved or, if necessary,
forwarded to one of the electronics
news groups.
Other hobbies that are supported
include radio controlled models and
SC
model trains.
http://www.hutch.com.au/~oztech/oztech.htm
Oztechnics homepage
http://motserv.indirect.com/
Mototola homepage
http://www.mot.com
Motorola corporate homepage
http://www.ee.ualberta.ca/html/
cookbook.html
Electronic cookbook Archive
Related sites
As well as the resources described
above, there are many web sits and
news groups that deal with the electronics aspects of various hobbies.
Model rocketry, a hobby of mine, is
Fig.7: Adobe’s Acrobat reader is being used here to display the datasheet on the
National Semiconductor LMX2216 0.1-2GHz low noise amplifier/mixer.
August 1996 7
CIRCUIT NOTEBOOK
Interesting circuit ideas which we have checked but not built and tested. Contributions from
readers are welcome and will be paid for at standard rates.
Tone burst source for
loudspeaker testing
Testing loudspeakers for maximum
power handling can be difficult if only
a continuous tone is available. This
is because the sound level is likely
to be excessive before the distortion
8 Silicon Chip
becomes obvious. Also it is not a
realistic test since music sources are
rarely continuous tones.
A tone burst signal can provide
a more valid test since it produces
a high level signal for a small time,
consistent with typical program material. The tone burst source presented
here will produce a 100Hz tone for
200ms every 1.4 seconds. The output
level is adjustable and connects to a
power amplifier in order to drive the
loudspeaker into overload.
Measuring the power level at overload is made simple by the addition
of a sampling circuit which monitors
Cordless telephone ring tone booster
The circuit uses a pickup coil to detect variations in the magnetic
field generated in the vicinity of the ringer. The view at right shows
the suggested mounting arrangement for the handpiece.
Many cordless telephones have a very feeble ring
tone, especially if one is challenged in the hearing
department. While the audio output may be weak, a
significant magnetic field is generated in the vicinity
of the ringer and this can be used to trigger a booster.
This device uses a Murata Sound Element (Cat. AB3444 from Jaycar) as the sound generator. The pick-up
coil consists of 800-1000 turns of 34 B&S (0.16mm)
wire on a 20mm diameter former, 5mm wide and with
a 5mm centre. A slug from a discarded IF transformer
will enhance sensitivity but is not vital.
Diodes D1 and D2 and the capacitors form a voltage
doubler to turn on Q1 when a ring signal is detected.
This, in turn, supplies positive bias to Q2, enabling the
sound element.
The circuit draws negligible standing current and
only 5mA when ringing so that battery life should approximate shelf life.
The hard part of construction is the support for the
handpiece (which will vary from model to model) to
hold the coil close to the ringer. The sketch shows one
possibility. The battery and PC board are mounted in
the hollowed out base.
A. March,
North Turramurra, NSW. ($30)
the signal during the tone burst. The
resulting signal is output as a DC voltage which can be read using a standard
multimeter. The power delivered to
the loudspeaker is simply calculated
by squaring the measured DC voltage
and dividing by the impedance of the
loudspeaker (2, 4 or 8Ω).
The circuit is based on the IHF Tone
Burst Source published in July 1988.
A 200Hz signal is produced by 555
timer IC1 which is divided by two in
D-flipflop IC3a. This gives a 100Hz
square wave. IC4a is connected as a
bandpass filter to reject signals other
than the 100Hz fundamental and the
output is thus a sinewave. VR2 adjusts
the signal level applied to IC4b.
Normally, with switch IC6a closed,
there is very little 100Hz signal at the
pin 1 output of IC4b. When the switch
opens, IC4b has a gain of 2.2 and the
signal level is high.
IC2 sets the burst rate and duration.
The pin 3 output controls the data input
of D-flipflop IC3b. Its high output time
is set by the 180kΩ and 27kΩ resistors
plus the 10µF capacitor. The low output
time is set by the 27kΩ resistor. Each time
IC3b is clocked by the 200Hz signal, the
Q output follows the data input which
in turn controls switch IC6a.
The amplifier output is monitored
and rectified by diode D6. The signal
is then divided by the attenuator set
by the resistors at switch S4. IC6b
allows the signal to pass to a 10µF
capacitor during the burst period
and it is controlled by the Q-bar output of IC3b. IC5 buffers the voltage
applied to the 10µF capacitor and
holds the voltage level when the
tone burst is off.
Power is derived from a 15V
transformer. D1-D4 rectify the AC,
while a 7815 regulator provides the
circuit with a 15V supply.
SILICON CHIP.
August 1996 9
SILICON
CHIP
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which is now out of date and the advertiser
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more than likely that it contained advertising
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ELECTRONIC
Clever IC
provides
rapid
turn-on
STARTER
For fluorescent lamps
Do your fluorescent lamps go blinkety-blink blink blink
when you turn them on? Or do they flash on to blind you and
then plunge you into darkness? Solve these problems with
this new electronic starter which gives a rapid start every
time. It fits in a standard starter case so the lamp wiring does
not have to be altered.
By JOHN CLARKE
L
ET’S FACE IT, fluorescent lights are bright and effi-
cient but they can be very annoying when they
don’t start as soon as you switch them on. This “blink
blink blink - nothing - flash - Ah! it’s on!” sequence can be
particularly frustrating if you need to leave your warm bed
on a cold night for a “comfort break”. Fluorescent lamps
are much harder to start when the temperature is low
which adds to the problem.
What can be really frustrating if you have
a cantankerous fluorescent lamp is that
changing to a new starter or even
a new tube may not help much.
Modern slimline 18W and 36W
tubes are hard to start, even when
new, and they are a real problem
if they are used in a batten fitting
intended for older style 20W or
40W tubes.
Up till now, there has been
no solution to this problem but
Philips has just released a surface
mount 8-pin chip which appears
to be a real ripper. Designated
the UBA2000T, it is specifically
designed to start slimline “TL” tubes
and incorporates features which overcome all the disadvantages of conventional
14 Silicon Chip
“glow switch” fluorescent starters.
Before delving into the operation of the electronic starter
we need to see how a fluorescent lamp circuit works and
why the conventional starter it has its disadvantages. So
let’s refer to Fig.1.
A fluorescent lamp is connected to the 240VAC mains
supply via a “ballast” which is an iron cored inductor.
In more detail, the current from the 50Hz
mains passes through one of the tube
filaments, then through the starter,
through the other filament and
then via the ballast. The starter,
as its name implies, gets it all
going.
If you pull a conventional
starter apart, and you will if
you build this project, you
will find that it contains what
looks like a conventional miniature neon lamp connected
in parallel with a high voltage
capacitor, typically .005µF 2kV
ceramic disc. This very simple
construction has quite a complex
function. Similarly, the fluorescent
tube itself looks very simple but there
is more to it than meets the eye.
A fluorescent tube is coated with
a phosphor on the inside of the glass
and it contains a minute quantity of
mercury and a mixture of inert gases.
As well, it has a filament heater at
each end.
This made with triple coiled tungsten wire and coated with an emissive
material such as barium or strontium
oxide. When power is first applied to
the circuit of Fig.1, current is passed
through the two filaments to raise them
to red heat and this causes them to
emit electrons, just like the filament
in a radio valve. The electrons rapidly
disperse along the tube so that when a
high voltage is applied to the tube, an
electric discharge can occur through
the inert gases.
Once this discharge starts, the
mercury in the tube is vaporised and
it begins to emit ultraviolet light. The
ultraviolet causes the tube phosphors
to fluoresce and so visible light is
produced.
The job of the starter is twofold.
First, it has to let current pass through
the filaments so they can heat up and
emit electrons. Then after a short delay, the starter interrupts the current
Fig.1: the circuit a
conventional fluorescent
lamp with a glow switch
starter. The starter enables
filament current to flow
at switch-on and it opens
after a short delay. The
back-EMF then generated
by the ballast inductor
then fires the tube. That’s
the theory anyway.
to the filaments. Since the ballast
inductor is also in series with the
filaments, this sudden interruption
of current causes it to produce a brief
high voltage spike. This high voltage is
applied to the tube to cause the electric
discharge referred to above. If all goes
well, the tube lights up and then the
starter is effectively out of circuit.
Clearly though, while the glass
tube in the fluorescent starter might
just look like a largish neon lamp, it
is more than that. The starter has two
contacts, one of which is bimetallic.
When voltage is first applied to the
circuit of Fig.1, the inert gas inside the
starter ionises and a small amount of
current flows. This heats up the interior of the starter and so the bimetallic
contact bends over slightly to meet its
mate and so current can flow through
the two filaments and the ballast.
Meanwhile the interior of the starter
cools down, the bime
tallic contact
opens the circuit, the filament current
stops and the ballast fires the tube. If
all goes well, that is. Generally though,
the starter has to make several tries
before the fluorescent tube fires properly and that leads to the blink, blink
problem that we all know and hate.
Features
•
•
•
Fig.2: functional diagram of the UBA2000T TL lamp starter. It counts the cycles
of the 50Hz supply to give a precise filament heating time and it also monitors
filament current to ensure that the lamp has the best chance of starting.
•
•
•
Starts 18 and 36W slimline
fluorescent tubes
Compatible with standard fluorescent starters
Fast start without excessive
flicker
Precise preheat time
Minimal radio interference
Timeout if lamp fails to fire
August 1996 15
Fig.3: the UBA2000T lamp
starter IC (IC1) switches
a 1000V Mosfet (Q1) to reliably
start slimline and conventional
fluorescent tubes. The IC
repeats the start sequence up
to six times, after which the
Mosfet is turned off as a safety
measure.
4x1N4007
So why is the capacitor inside the
starter? One reason is that it helps
prevent arcing across the contacts as
they open. The other is that it helps
reduce the radio interference both
from the starting operation and from
the electric discharge inside the fluorescent tube.
These conventional starters are very
simple to manufacture but they have
a number of drawbacks. First, the pre
heat time is set by the thermal lag of
the bimetallic contact. This is the time
it takes the contact to cool and reopen
and it can vary depending on ambient
temperature and manufacturing tolerances. In some cases the preheat time
will not be enough to allow the filaments to warm up sufficiently to fire
the tube. Naturally, this problem gets
worse as the starter and fluorescent
tube get older.
A more serious problem is that when
the starter contact opens, the induced
voltage from the ballast may not be sufficient to fire the tube. This is because
the bimetallic contact can open at any
time within the mains cycle and the
ballast current may be very low when
this happens. So that is why even a
new starter may need several tries to
fire the fluoro tube.
PARTS LIST
1 PC board coded 10308961, 17
x 28mm
1 fluorescent starter container
and lid with terminals (see
text)
1 12mm diameter x 12mm long
piece of heatshrink tubing
Semiconductors
1 UBA2000T TL-lamp starter
(IC1) (Philips)
1 TO-220 1000V Mosfet,
BUK456-1000B, STP3N-100
(Q1)
4 1N4007 1000V rectifier diodes
(D1-D4)
Capacitors
1 3.3µF 63VW PC electrolytic
1 .0056µF 2kV ceramic
Resistors (0.25W, 1%)
1 1MΩ
1 100kΩ 500V MAX (Multicorp)
1 62kΩ
Thirdly, there is no provision to
stop the starter sequence if the lamp
fails to start. This repetitive starting
can eventually burn out the ballast
Fig.4: this diagram illustrates the starting sequence of the UBA2000T.
16 Silicon Chip
due to overheating. Alternatively, if
the starter’s contacts weld up, the ballast will be burnt out and this means
an expensive repair. Generally, it is
cheaper to replace the whole lamp
fitting.
Clever chip
Our new electronic starter circuit is
shown in Fig.2. It plugs in directly to
the starter socket on a fluorescent lamp
fitting. As well as using the Philips
UBA2000T lamp starter chip, it has
a 1000V Mosfet, a bridge rectifier and
a few resistors and capacitors. While
the UBA2000T is a teensy little chip,
it has quite a lot of circuitry inside it,
as indicated by the functional diagram
of Fig.2.
Looking at Fig.2, the UBA2000T has
Vin and Vsense pins which monitor the
mains voltage and filament current,
respective
ly. By monitoring Vin the
UBA2000T knows whether the tube
is ignited or not; the voltage level
is lower once the tube is ignited. By
monitoring the filament current, the
UBA
2000T can fire the tube at the
optimum time.
Pin 3 drives the gate of a 1000V
Mosfet which is used to switch the
filament current on and off. The Mosfet
is not switched on if the Vcc supply is
too low (below 40-49V) or the current
through the filaments is excessive
(above 2.2A peak).
Fig.4 shows the typical start sequence waveform. When power is first
applied to the circuit, the capacitor
at the Vcc pin is charged through the
internal switch S1. When Vcc reaches
the start voltage, (Vcc(sl)) and when
the mains voltage is at its peak value,
the Mosfet will be turned on.
The UBA2000T now counts the
mains cycles until 1.52 seconds (ie,
76 cycles at 50Hz) has elapsed. Also
during this time the capacitor at the
Vcc pin discharges. The Mosfet is
switched off provided the current
through the internal sense resistor is
greater than 285mA. This allows the
ballast inductor to produce sufficient
voltage to fire the tube. Typically,
this firing voltage will be somewhere
between 700 and 800V!
If the fluorescent tube does not fire,
the UBA2000T tries again, as shown
in Fig.4. It must first recharge its own
supply capacitor at pin 6 (Vcc) and
then filament current is applied again.
This second preheat period is set to
0.64s since the filaments are already
assumed to be warm.
After the tube fires, the peak voltage
across it will typically be about 100V
which is considerably lower than the
mains voltage peak (around 340V)
and so the Vcc(sl) threshold for the
UBA2000T can no longer be reached.
The Mosfet is therefore held off and
the starter circuit is effectively out of
action until the mains voltage is turned
off and reapplied.
The UBA2000T will repeat the
start sequence six times after which
the Mosfet will be turned off. This
is a very good safety feature since it
prevents the ballast inductor from
being burned out. As a further safety
feature, the Mosfet will be turned off
if the sensed preheat current exceeds
2.2 amps peak.
Circuit description
The circuit of Fig.3 shows how the
UBA2000T is used in practice. Diodes
D1-D4 are connected in a bridge to
rectify the mains voltage. This applies
the correct voltage polarity to both
IC1 and Q1.
The 100kΩ and 62kΩ resistors
across this rectified mains supply
divide the voltage down for the pin 4
input and limit the charge current to
the 3.3µF Vcc capacitor at pin 6.
The 1MΩ resistor between pin 6 and
the gate of Q1 provides a small pullup
This photo shows the copper side of
the assembled PC board. The surface
mount IC means that you will need a
fine-tipped soldering iron to mount it
in place.
Fig.5: the parts layout diagrams
for both sides of the PC board.
Note that the four diodes and
the 100kΩ resistor are mounted
underneath the .0056µF ceramic
capacitor.
Only three parts are mounted on this
side, the main one being the 1000V
Mosfet. This should be sleeved with
heatshrink tubing before the board is
installed inside the starter case.
current to keep the Mosfet gate high
once it is triggered by a pulse from
pin 3. The gate is switched off when
pin 3 goes low. Note that the source
electrode of the Mosfet is connected
to pin 1 so that its current (ie, the
filament current) is sensed by the
internal 26mΩ resistor between pins
1 and 2 of IC1.
Capacitor C1 is included to suppress
radio frequency interference caused by
discharge in the tube.
Note that all components are rated
for the high voltages involved. The
3.3µF 63VW capacitor can have up to
49V across it, while the voltage across
the tube at the instant Q1 is switched
off can be 800V or more. Consequently,
C1 has a 2kV rating while Q1 and D1D4 have a voltage rating of 1kV. The
100kΩ resistor must have a minimum
rating of 500V.
Construction
Fig.6: this is the full-size etching
pattern for the PC board. Check
your board for etching defects by
comparing it with this pattern
before installing any parts.
The electronic starter is constructed
on a PC board coded 10308961 and
measuring 17 x 28mm. This is designed to be a snug fit inside a standard
fluorescent starter container. Even so
we had to mount components on both
sides of the board and in some cases
RESISTOR COLOUR CODES
❏
❏
❏
❏
No.
1
1
1
Value
1MΩ
100kΩ
62kΩ
4-Band Code (1%)
brown black green brown
brown black yellow brown
blue red orange brown
5-Band Code (1%)
brown black black yellow brown
brown black black orange brown
blue red black red brown
August 1996 17
The electronic
fluorescent starter
is mounted inside
a dud fluorescent
lamp starter case.
It will rapidly start
slimline (25mm)
and conventional
(38mm) fluorescent
tubes without
flashing.
they lie on top of each other, as you
will see from the diagram of Fig.5 and
the photos.
To put this PC board together you
will need either a very keen pair of
eyes or better still, a pair of close-up
specs or a mag-lamp. IC1 is a surface
mount IC so you will need a finetipped soldering iron since the IC’s legs
are spaced only 1.27mm (.050") apart.
Note that since IC1 is a surface-mount type, it is mounted on
the copper side of the board. Other
components on the copper side are
the four diodes, C1 and the 100kΩ and
62kΩ resistors.
Check that the PC board is correct
by comparing it with the published
pattern. Correct any shorted or broken
tracks at this stage. Before soldering
anything to the board we suggest that
you pre-tin the copper tracks for the
IC pins. Then place the IC in position
making sure it is oriented correctly.
How do you do that?
We did say you will need very good
vision. Notice that one end of the IC
is chamfered along one side; pin 1 is
at the top, if you hold the IC with the
chamfered edge at left. This can be
seen in Fig.5.
Once you have the tracks for IC1
tinned and it is in position, solder each
pin quickly with just enough heat to
melt the solder on the PC board. Then
check each solder connection is good
by measuring the resistance between
each pin and the PC track.
On the component side of the PC
board side insert and solder in the
1MΩ resistor and the 3.3µF capacitor, taking care with polarity. The
capacitor should lie over the 1MΩ
resistor. Keep a few mm of lead length
between the PC board and capacitor so
that it lies more or less parallel with
the board. Cut the leads short on the
copper side after soldering.
Now install the remaining parts on
the copper side. Make sure that the
diodes are oriented as shown and cut
the leads flush with the PC board side.
Now place the .0056µF capacitor over
the diodes and bend its leads so that
the capacitor body can lie parallel with
the board. Solder this in place.
Next, mount the Mosfet on the
component side, with its leads bent at
right angles so that it lies parallel to
and close to the board. It is oriented
so that the metal tab faces away from
the top of the board. Solder and trim
its leads. Lastly, fit a 12mm length of
12mm diameter heatshrink tubing over
the Mosfet to completely insulate it.
Starter container
You will need to disassemble a
starter for its case and lid with terminals. Use a small screwdriver to
carefully prise the Bakelite lid from the
cylindrical container. You will need
to gradually work around the whole
circumference of the container with
the screwdriver until the baseplate
can be removed.
Withdraw the lid and components
and cut the wires close to the capacitor
body. These leads are then attached to
the PC board of your new electronic
starter. Cut the starter tube wires close
to the baseplate lug. The capacitor
leads can now be inserted into the
PC board from the PC board side and
soldered in place.
Next, carefully inspect the PC board
assembly for any solder dags or splashes or pigtails which are too long. This
aspect is most important when you
consider the peak voltages which can
occur between the leads to the Mosfet
and diodes. None of your soldering
should diminish the gaps between
conductors of the bare board.
When you are satisfied that all aspects of the soldering and assembly are
correct, insert the PC board and starter
lid assembly into the container and
clip in place. Note that the components
may need to be pressed closer to the
PC board if the fit is too tight.
Finally, test your new electronic
starter in a fluorescent light fitting. The
tube should initially glow orange at the
filaments, then glow white at the tube
ends and then light up fully, usually
SC
after the first attempt.
Especially For Model
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August 1996 19
ails
t
e
D
t
i
u
rc
Pt.2: Ci
Build a VGA
digital oscilloscope
One of the real attractions of this digital scope,
based on a VGA monitor, is that you don’t have
to peer at the display – it is large, bright and the
different coloured traces and graticule make
it easy to interpret what’s happening. In this
article we discuss the circuit details.
By JOHN CLARKE
To fully understand the discussion,
it will be a help if you can refer to the
block diagram presented on page 28
of last month’s issue.
The circuit for the VGA Oscilloscope has been split into two sections.
The main section is Fig.1, on pages 66
and 67, comprises most of the circuit
20 Silicon Chip
while the timebase circuit, Fig.3, is
shown on page 69.
28 ICs are used in the entire circuit.
However, as we shall see, some of the
circuit is repetitive (for the CH1 and
CH2 inputs) and much of it involves
timers and counters.
Looking at the top righthand corner
of the two-page circuit, S1 is the AC/
DC coupling switch for the Channel
1 input. S1 also has a GND setting to
allow the trace to be positioned on the
graticule as a ground (zero) reference
point.
Following S1, the input signal
passes via the attenuator switch S2.
This is essentially a string of resistors, with each one bypassed by a
capacitor to improve high frequency
response. Trimmer capac
itor VC1
allows adjustment of the frequency
compensation.
After the attenuator, the signal is
applied to the gate of JFET Q1 which
acts as a high impedance buffer. Its
gate is protected from excessive signal
excursion by two back-to-back LEDs.
These begin to conduct for signals in
PARTS LIST
1 plastic case, 262 x 189 x
84mm, with metal panel
1 front panel label, 252 x 76mm
1 PC board, code 04307961,
252 x 75mm
1 PC board, code 04307962,
213 x 142mm
1 PC board, code 04307963,
252 x 75mm
2 PC boards, code 04307964,
20 x 32mm
3 2P3W slider switches
(S1,S3,S11)
3 1-pole 12-way rotary switches
(S2,S4,S5)
5 SPDT toggle switches
(S6,S7,S8,S9,S12)
1 SPDT centre-off switch (S10)
2 10kΩ horizontal mount
trimpots (VR1,VR3)
1 5kΩ horizontal trimpot (VR6)
2 500Ω linear pots (VR2,VR3)
1 5kΩ linear pot (VR5)
2 2-47pF miniature trim
capacitors (VC1,VC2)
1 9-68pF miniature trim
capacitor (VC3)
1 4MHz parallel resonant crystal
(X1)
1 15-pin VGA line socket and
lead
1 cable clamp
1 5mm rubber grommet
1 DC panel socket
2 BNC panel sockets
5 3mm LEDs (LEDs 1-5)
3 15mm OD black knobs
3 18mm OD knobs (1 green, 1
blue, 1 red)
92 PC stakes
2 8-way pin headers
1 1.8m length of 0.8mm tinned
copper wire
1 150mm length of shielded
cable
1 800mm length of 4-way
rainbow cable
1 400mm length of red hookup
wire
1 400mm length of green
hookup wire
1 400mm length of blue hookup
wire
1 400mm length of yellow
hookup wire
1 400mm length of black hookup
wire
3 3mm diameter x 6mm machine
screws and nuts
excess of ±1.8V peak and are there
mainly to cater for the situation where
the input attenuator is set too low for
the size of the signal. Normally, if the
attenuator is correctly set, the signal
at the gate of Q1 will not exceed about
±200mV peak.
IC1 and IC2 invert and amplify the
signal by about 25 times to produce
sufficient level for the following A-D
converter which requires 5V for full
conversion. VR2 controls the DC output offset of IC1 and IC2 and thereby
has the effect of shifting the signal
Semiconductors
4 CA3140 op amps (IC1,IC2,
IC7,IC8)
2 ADC0820CCN 8-bit A-D
converters (IC3,IC9)
2 MCM6206DJ20 20ns 8-bit
RAMs (IC4,IC10)
4 74HC85 4-bit magnitude
comparators (IC5,IC6,IC11,
IC12)
4 7555,TLC555CN CMOS timers
(IC13,IC20,IC22,IC28)
4 74HC74 dual D-flipflops
(IC14,IC19,IC26,IC27)
1 74HC86 quad EXOR gate
(IC15)
1 74HC4053 analog CMOS
switch (IC16)
2 74HC193 4-bit presettable
counters (IC17,IC18)
1 LM319 dual comparator
(IC21)
2 74HC00 quad NAND gates
(IC23,IC29)
2 74HC4040 binary counters
(IC24,IC25)
1 7812 12V regulator (REG1)
1 7805 5V regulator (REG2)
2 2N5484 JFETs (Q1,Q2)
3 BC338 NPN transistors
(Q3,Q6,Q9)
2 BC548 NPN transistors
(Q4,Q7)
2 BF199 NPN RF transistors
(Q5,Q8)
21 1N914 signal diodes (D1D16,D21-D25)
4 1N4004 diodes (D17-D20)
5 3mm red LEDs (LED1-5)
Capacitors
1 1000µF 16VW PC electrolytic
1 33µF 16VW PC electrolytic
15 10µF 16VW PC electrolytic
1 6.8µF 16VW PC electrolytic
1 1µF 16VW PC electrolytic
2 0.22µF MKT polyester
14 0.1µF MKT polyester
1 .047µF MKT polyester
4 .0039µF MKT polyester
2 .0015µF MKT polyester
5 .001µF MKT polyester
2 680pF MKT polyester or ceramic
1 560pF MKT polyester or
ceramic
1 470pF MKT polyester or
polystyrene
2 390pF ceramic
2 150pF ceramic
2 82pF ceramic
3 47pF ceramic
2 22pF ceramic
3 3-60pF trimmer capacitors
(VC1-VC3)
Resistors (0.25W, 1%)
1 10MΩ
1 20kΩ
1 3.9MΩ
2 12kΩ
1 2.2MΩ
6 10kΩ
1 820kΩ
3 7.5kΩ
2 510kΩ
1 6.8kΩ
1 390kΩ
1 3.9kΩ
2 240kΩ
1 3.3kΩ
1 220kΩ
5 2.7kΩ
1 150kΩ
11 2.2kΩ
2 130kΩ
2 1.8kΩ
3 100kΩ
1 1.5kΩ
1 82kΩ
7 1kΩ
2 75kΩ
2 330Ω
2 51kΩ
1 220Ω
2 47kΩ
1 120Ω
3 39kΩ
3 75Ω
2 27kΩ
Miscellaneous
Solder, four self-tapping screws,
cable ties.
Fig.1 (following page): the main
circuit section for the VGA
Oscilloscope. This comprises the
input circuitry, A-D converters
(IC3 & IC9), memory storage
devices (IC4 & IC10) and the
oscilloscope timebase circuitry
(IC13-15, IC17 & IC18).
August 1996 21
22 Silicon Chip
August 1996 23
Fig.2: these
oscilloscope
waveforms show
the timing for the
record sequence.
The top trace is
the read/write
input of the A-D
converters, while
the middle trace
is the enable input
for the memory.
The lower trace is
the clock input to
counter IC17.
trace up or down the VGA screen. VR1,
in the feedback loop for IC2, changes
the gain for the vertical calibration
function.
Note that any change in the setting
of VR2 will affect the overall gain of
IC1 and IC2 since this is part of the
gain setting resistance.
However, the range over which the
potentiometer is adjusted to set the
waveform fully up or fully down on
the screen is only a small percentage
change compared to the overall resistance value. As a result, the gain
change in not perceivable on the
screen.
IC1 and IC2 are powered from 12V
in order to ensure an output swing
capability of more than 5V, while
diodes D1-D3 clamp IC2’s output to
prevent overload in the following A-D
converter.
IC3 is an 8-bit high speed A-D converter with an inbuilt sample and hold
feature. It has a 4-bit flash converter
which uses 32 comparators to speed
up conversion and can convert an
analog signal to an 8-bit digital code
in a maximum of 1.5µs.
AD conversion is started by a low
on the WR-bar input at pin 6 of IC3.
This must be low for at least 600ns
before going high and must remain
high for a minimum of 800ns before
the data is valid.
The data output lines are connected
to the RAM chip IC4. This is a 20ns
access time high speed memory containing 32K bytes. We have only used
256 bytes and although this may seem
wasteful, its selection was based on
the high speed and cost. Paradoxically,
24 Silicon Chip
larger memory can be less expensive
than the less popular smaller RAM
chips.
High speed RAM is paramount for
this application. Remember that when
the memory is called to cycle through
each location when displaying the
stored waveform on the screen, the
allotted time per memory location is
only 125ns (4MHz rate or 250ns period
and 125ns per half cycle).
This means that there are 20ns devoted to accessing the correct data and
105ns devoted to comparing this value
with the line counter. Any standard
120ns memory would be lost trying
to keep up this pace.
Channel 2 signals
The signal process for channel 2 is
identical to that described above, the
path being via attenuator switch S4,
buffer Q2 and amplifiers IC7 and IC8.
A-D conversion is in IC9 and the data
is stored in RAM chip IC10.
IC17 and IC18, which are synchronous 4-bit preloadable counters, drive
the address lines of IC4 & IC10. The
clock input at pin 5 of IC17 (which is
also the A0 input for IC4 and IC10) is
from IC16 at pin 4.
When the oscilloscope is in display
mode the clock signal comes from the
MAGnification selection at S11a via
pin 5 of IC16 and is fed through to pin
4. When in the record mode, the clock
is from IC15c at pin 3 of IC16. This
indirectly obtains a clock signal from
the timebase oscillator, IC13.
Triggering
The outputs of IC2 and IC8 connect
to switch S6 and this selects the source
of triggering from channel 1 or channel
2. Comparators IC21a and IC21b take
the signal from S6 to generate the
trigger signal.
IC21a generates trigger signals for
positive-going signals while IC21b
acts as an inverter to generate trigger
signals for negative-going inputs.
The trigger threshold (level) is set by
VR5. Positive or negative triggering is
selected by switch S7.
The comparator output selected
by S7 triggers IC22 which is a 7555
timer set up as a one shot. When triggered, its output at pin 3 goes high
and remains high until reset by the
update oscillator IC20. This occurs
when S8 is in the realtime position
but IC22 remains set if left in the
store position.
IC20 is another 7555 timer which
operates as a free running oscillator.
It charges the selected capacitor at pin
2 and 6 via a 6.8kΩ resistor and diode
D7 and discharges it via the 150kΩ
resistor.
With this setup, its pin 3 output is
high for a short time (to reset IC22) and
low for a relatively long time to allow
triggering from IC21.
Flipflop IC19b is triggered either
by IC22 or IC20, depending on the
setting of switch S9. In the free run
position of S9, the display is updated
at a regular interval set by the frequency of IC20. This means that the
display will not be locked (ie, it will
be moving) since a different part of
the waveform will be stored at each
trigger point.
The triggered selection for S9
provides a static display since the
waveform is stored at the same point
in the waveform each time and only
when the pin 3 output of IC20 is high.
IC19b is reset when power is first
applied, due to the 10µF capacitor at
the cathode of D11 being discharged.
This pulls the CLR input (pin 13) low
to reset the Q output low and the Q-bar
output high. When IC19b is triggered,
Fig.3 (right): the VGA timebase
circuit. NAND gate IC23a and X1
form a master crystal oscillator, while
binary counters IC24 & IC25 provide
the 8-bit data signals for IC5, IC6,
IC11 & IC12 (on Fig.1).
August 1996 25
Fig.4: these
oscilloscope
waveforms show
the line sync
pulses (top) and
the frame sync
pulses (bottom ).
The centre trace
is actually a
horizontal line for
the graticule.
the Q output at pin 9 goes high and
operates the three switches inside
IC16 via the A, B and C inputs. This
switches pin 4 to pin 3, pin 15 to pin
1 and pin 14 to pin 13.
At the same time, the low output
at pin 8 of IC19b (Q-bar) selects the
ADC chips IC3 and IC9 (via their the
CS inputs) and places IC4 and IC10
(RAM) in the write mode.
The low Q-bar output from pin 8
of IC19b also clears IC19a, via the
560pF capacitor connected to pin 1.
This causes the Q-bar output of IC19a
to go high.
Before this happens, the previously
low Q-bar output of IC19a presets IC17
and IC18, via IC16. Diode D12 holds
the C preset input (pin 10) of IC17
low and so both counters are preset
to 0000 0000. An RC delay from the
Q-bar output of IC19a to pin 13 of IC16
is used to extend the preset time for
IC17 and IC18.
The above sequence sets the circuit
in the record mode. Timebase oscillator IC13 now controls the read/write
inputs of A-D converters IC3 and IC9.
Fig.2 shows a screen printout of
oscilloscope waveforms of the timing
for the record sequence. The top trace
is the read/write input of the A-D
converters.
When low, the A-D converter samples the data and flash converts the
four most signifi
cant digits. 800ns
after the rising edge of the read/write
input, data from the A-D converter
is valid.
The middle trace of the oscilloscope
waveform is the enable input for the
26 Silicon Chip
memory. It is derived from the time
base and passes through EXOR gate
IC15a (IC15 is near the lower righthand
corner of the circuit, Fig.1).
IC15a has its pin 1 input directly
connected to the timebase, while pin
2 is connected via an RC delay. Whe
never the input to IC15a changes, pin
3 goes high for the delay period to
disable the memory. This prevents
any false data from the A-D converter
being applied to the data inputs of
the memory.
The lower trace on Fig.2 is the
clock input to counter IC17 which is
a divide-by-two timebase signal. The
timebase clocks IC14, which is connected as a toggle flipflop. Its output
is passed through two EXOR gates,
IC15b & IC15c, wired as non-inverters to introduce a small amount of
delay between the positive edge of the
timebase and the change in the clock
signal for IC17.
This ensures that the memory is disabled via IC15a (middle trace) before
the address is changed.
When counters IC17 & IC18 have
reached a count of 256, the QD output
at pin 7 of IC18 goes high and clocks
D-flipflop, IC19a. This produces a low
at the Q-bar (pin 6) output of IC19a and
this presets IC17 & IC18.
A low pulse to the CLR input
of IC19b via the 680pF capacitor
sets the Q output of IC19b low and
Q-bar high. The high Q-bar output deselects A-D converters IC3
and IC9 and switches the IC4 and IC10
memories to read mode via the Writebar inputs at pin 27.
The low Q output switches IC16
so that the clock input of IC17 and
address line 0 of IC4 are controlled
by the 4MHz to 1MHz inputs at pin 5.
These inputs come from the timebase
circuit (see Fig.3). Also the memories
are permanently enabled via the now
low E-bar inputs caused by pin 15
of IC16 connecting to the low pin 2.
Counters IC17 and IC18 are now preset
by the line sync signal now present
at pin 14.
In addition, the low Q-bar output
of IC19b clears IC19a via the 560pF
capacitor connected to pin 1. Its Q-bar
is high and so diode D12 connecting to
the C input of IC17 is reverse biased.
This input is pulled high via the 10kΩ
resistor when S11b is in position 1.
Preloading of IC17 now sets it to an
initial count of 8.
This may appear unusual, however
it is used to move the oscilloscope
trace along the screen so that the trigger point is exactly in line with the far
left graticule vertical line.
Magnitude comparators
IC5 and IC6 (top righthand corner
of Fig.1) are digital magnitude compar
ators with “less than”, “greater
than” and “equal-to” outputs. For our
application we only use the “equalto” output, pin 6, which turns on
the display when the data from IC4
(channel 1 RAM) is identical to the
line count from the VGA timebase
circuitry.
Pin 6 drives the base of transistor
Q3 via a 2.2kΩ resistor. This is level-clamped using diode D13 and the
1.8kΩ resistor. The emitter follower
configuration of Q3 drives the video
input (green) of the VGA monitor via
a 75Ω resistor. Thus the channel 1
trace is green.
Transistors Q4 and Q5 are for blanking the display when updating the
A-D conversion and during the line
sync pulse respectively. This prevents
the trace from producing an unusual
display or overscanning.
Q6, Q7 and Q8 operate in the same
manner as above for the channel 2
trace; ie, Q6 drives the red gun of the
VGA monitor, while Q7 & Q8 are for
blanking. Q6 is driven by pin 6 of IC12.
IC11 & IC12 are the digital magnitude
comparators for channel 2.
Power
Power for the circuit is derived from
a 12VAC plugpack which is rectified
using D17-D20 and filtered with a
1000µF capacitor. REG1 and REG2
provide the +12V and +5V supplies
for the circuit.
The 10µF capacitors at the output
of each regulator prevent instability.
There are also a number of 10µF and
0.1µF decoupling capacitors across
the supply rails.
VGA timebase
The VGA timebase circuit is shown
in Fig.3. NAND gate IC23a is the master
crystal oscillator operating at 4MHz.
A 10MΩ resistor between pins 11 and
12 biases the inverter into the linear
mode. Crystal X1 oscillates across the
inverter pins with the 22pF capacitors
providing loading to prevent overtone
oscillation.
IC23b inverts the clock signal and
both this signal and the pin 11 output
from IC23a is applied to the Data (pin
2) and the Preset (pin 4) inputs of flipflop IC26b. IC26b is a D-flipflop and
the inverted level at the Data input is
clocked through to the Q-bar output
on the positive edge of the CK input
at pin 3.
When the preset is low, the Q-bar
output is set low.
Graticule generation
IC24 is a binary counter with outputs from Q1-Q12. These are advanced
on the negative transition of the clock
input at pin 10. Its Q4 output runs at
250kHz and this is inverted with IC29a
to clock IC26b.
When Q4 goes low and when the
Preset input of IC26b is high, the Q-bar
output of IC26b goes high. As soon as
the Preset goes low again after 125ns
the Q-bar output goes low again. This
output produces a vertical graticule
line signal 125ns wide and is repeated
at a 250kHz rate or every 4µs.
This means that we have 8 vertical
graticule lines in the allotted 32µs for
each line.
The vertical line signal drives buffer
transistor Q9 via a 1kΩ resistor. The
three series diodes limit the base drive
to 1.8V and the emitter to 1.2V.
The line sync pulses are derived
using IC26a which works in the same
manner as IC26b.
When the Preset input at pin 10 is
low, the Q output at pin 9 goes high
when Q7 of IC24 goes low. The result
is a 2µs (set by the Q3 output of IC24)
low-going pulse at the Q output of
IC26a.
This occurs every 32µs as set by
the Q7 output of IC24. Note that the
use of IC23c to NAND the Q3 and Q4
outputs of IC24 before being applied
to the Preset input of IC26a essentially shifts the line sync pulse so
that it occurs before the first vertical
graticule line.
IC25 is a second binary counter
to give us the requisite Q13-Q16
outputs. Note also that Q9-Q16 are
the line counter outputs used for
comparators IC5 and IC6 and IC11
and IC12.
Similarly, the 4MHz, 2MHz and
1MHz outputs are used to clock the
memories when in the playback mode,
as selected by switch S11a.
Frame sync pulses
Frame sync pulses are derived in a
similar way to the vertical graticule
line and line sync pulses. Q16 from
IC25 provides a 61Hz signal, while
Q9 from IC24 gives a 64µs pulse
width.
The oscilloscope waveforms in
Fig.4 show the line sync pulses (top
trace) and the frame sync pulse (bottom trace).
The centre trace is actually a horizontal line for the graticule. Note that
it occupies an entire line height from
one line sync pulse to another.
IC28 is triggered by the Q13 output
which occurs at eight times the frame
frequency. This gives us a possible
eight horizontal graticule lines.
Unfortunately, this number does
not result in a graticule in the centre
of the screen. And a central graticule
line is a very desirable oscilloscope
feature.
In order to obtain this, timer IC28 is
used to delay the occurrence of each
line so that one will actually be in
the centre.
The .047µF capacitor along with the
3.9kΩ resistor and trimpot VR6 set the
delay at about 2048µs.
The horizontal line signal from
IC28 is inverted and clocked through
IC27b for a horizontal line signal
which is locked into the line sync
pulse.
The horizontal graticule line is
also buffered by Q9 before being
applied to the blue gun input of the
monitor.
This completes the circuit description. Next month we will present
the construction details of the VGA
SC
Oscilloscope.
TRANSFORMERS
• TOROIDAL
• CONVENTIONAL
• POWER • OUTPUT
• CURRENT • INVERTER
• PLUGPACKS
• CHOKES
STOCK RANGE TOROIDALS
BEST PRICES
APPROVED TO AS 3108-1990
SPECIALS DESIGNED & MADE
15VA to 7.5kVA
Tortech Pty Ltd
24/31 Wentworth St, Greenacre 2190
Phone (02) 642 6003 Fax (02) 642 6127
If you are seeing a
blank page here, it is
more than likely that it
contained advertising
which is now out of date
and the advertiser has
requested that the page
be removed to prevent
misunderstandings.
Please feel free to visit
the advertiser’s website:
www.latrobe.edu.au/
August 1996 27
SILICON
CHIP
If you are seeing a blank page here, it is
more than likely that it contained advertising
which is now out of date and the advertiser
has requested that the page be removed to
prevent misunderstandings.
Macservice Pty Ltd
SILICON
CHIP
If you are seeing a blank page here, it is
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prevent misunderstandings.
Macservice Pty Ltd
Rugged Mosfet
Audio Amplifier
Module
By LEO SIMPSON
Want a big powerful amplifier module based on
Mosfets? This one uses eight plastic Mosfets to
deliver just over 200 watts into an 8W load and a
whisker over 350 watts into a 4W load – just the
ticket for heavy duty amplification.
For many audio enthusiasts, Mosfets rule supreme and Hitachi Mosfets
are the best there are. But in the last
few years, Exicon, a manufacturer
from England, has appeared on the
scene with a range of plastic power
Mosfets. This new design features
these plastic devices which are
rated at 20 amps, 200V and 125W.
Eight of these devices – ie, four Exicon ECX10P20 p-channel and four
ECX10N20 n-channel – are used in
this amplifier module.
As the graphs of Fig.1 & Fig.2
demonstrate, the amplifier module
will deliver just over 200 watts into
an 8Ω load or just over 350 watts into
a 4Ω load, at the onset of clipping.
The onset of clipping is where the
harmonic distortion graph suddenly
becomes almost vertical.
While we’re talking about performance graphs, we might as well refer
30 Silicon Chip
to a few more. Fig.3 shows the frequency response which is 0.7dB down at
10Hz and 20kHz. While it is just off
the graph, the -3dB point is at 54kHz.
Fig.4 shows the harmonic distortion versus frequency for the power
amplifier module when delivering
250 watts into a 4Ω load. Fig.5 shows
harmonic distortion versus frequency
at 150 watts into an 8Ω load. As these
graphs show, the performance is quite
respectable.
The amplifier module is also very
quiet, which is as it should be for any
modern design. We measured a signalto-noise ratio of 117dB unweighted
(22Hz to 22kHz) and 123dB A-weight
ed with respect to full power into an
8Ω load.
The PC board is designed so that
the eight Mosfet power devices are
mounted onto a heatsink angle bracket
which then mounts on a large finned
heatsink as part of the amplifier chas
sis. Our photos show only the heatsink
bracket. The amplifier must not be
operated without a larger heatsink
as it will rapidly overheat.
Circuit description
Fig.6 shows the circuit diagram.
This amplifier is unlike most direct-coupled circuits in that it has
three differential stages to give it high
open-loop gain before negative feedback is applied.
Two BC546 NPN transistors, Q4 &
Q5, form the differential input stage
and their operating current is set by
the constant current source, Q7.
The signals at the collectors of Q4
and Q5 are then fed into the voltage
gain stage which comprises Q1, Q2,
Q3, Q6, Q8, Q9 and associated components. This can best be described
as a “double differential pair with
TO
N
I
W
200
S;
8-OHM TO
IN
350W MS
4-OH
current mirror load”. This stage works
as follows. PNP transistors Q2 and Q3
form the first differential pair with R8
as the common emitter resistor. The
output of Q2, Q3 provide differential
drive to NPN transistors Q6 & Q8. The
collector load for these two transistors
is provided by the current mirror transistors Q1 & Q9.
The current mirror ensures equal
current sharing in the associated differential pair and thereby provides
high gain and good linearity.
Finally, we come to the power output stage which is the business end
of the amplifier; it employs the eight
Mosfets mentioned earlier. These
are connected as complementary
source-followers which means that
they behave in a similar way to emitter
followers – their voltage gain is a little
less than unity but they have oodles
of current gain. In effect, the Mosfets
act as a buffer stage for the amplifier,
transforming the voltage drive from
the earlier stages to a low impedance
output which can deliver a great deal
of power – 350 watts in fact!
The signal at the collectors of Q8
and Q9 (ignore VR1 for the moment)
is applied to the gates of the paralleled
Mosfets, via the 390Ω resistors. As the
signal rises towards the positive rail,
the top Mosfets (10N20’s) start to conduct, allowing current to flow to the
load. Conversely when the signal goes
towards the negative rail, the bottom
Mosfets (10P20’s) conduct, pulling
current out of the load.
Performance
Output power ......................... 200 watts into 8Ω; 350 watts into 4Ω
Frequency response .............. -0.7dB down at 10Hz and 20kHz (see Fig.3)
Input sensitivity ...................... 1.7V RMS (for 200 watts into 8Ω)
Harmonic distortion ............... less than .01% (see Figs.1 & 2)
Signal to noise ratio ���������� 117dB unweighted (22Hz to 22kHz); 123dB
A-weighted with respect to full power into 8Ω
Stability .................................. unconditional
August 1996 31
LEVEL(W)
AUDIO PRECISION SCTHD-W THD+N(%) vs measured
10
29 MAY 96 14:55:34
1
AUDIO PRECISION SCTHD-W THD+N(%) vs measured
10
LEVEL(W)
29 MAY 96 14:58:49
1
0.1
0.1
0.010
0.010
0.001
0.001
.0005
.0005
0.5
1
10
100
300
0.5
1
10
100
500
Fig.1: total harmonic distortion versus power into an 8Ω
load. Power at the onset of clipping is 212W.
Fig.2: total harmonic distortion versus power into a 4Ω
load. Power at the onset of clipping is 353W.
AUDIO PRECISION SCFREQRE AMPL(dBr) vs FREQ(Hz)
5.0000
AUDIO PRECISION SCTHD-HZ THD+N(%) vs FREQ(Hz)
5
28 MAY 96 11:10:30
29 MAY 96 15:11:42
4.0000
1
3.0000
2.0000
1.0000
0.1
0.0
-1.000
0.010
-2.000
-3.000
-4.000
0.001
-5.000
.0005
10
100
1k
10k
50k
Fig.3: frequency response of the amplifier. While it is just
off the graph, the upper -3dB point is at 54kHz.
AUDIO PRECISION SCTHD-HZ THD+N(%) vs FREQ(Hz)
5
29 MAY 96 15:02:38
1
0.1
0.010
0.001
.0005
20
100
1k
10k
20k
Fig.5: total harmonic distortion versus frequency at 150W
into an 8Ω load.
The 390Ω gate resistors are there to act as “stoppers” for
the Mosfets. They act in conjunction with the high gate
32 Silicon Chip
20
100
1k
10k
20k
Fig.4: total harmonic distortion versus frequency for the
amplifier module when delivering 250W into a 4Ω load.
capacitance of the Mosfets to reduce their gain at very
high frequencies. This prevents the tendency of Mosfets
to “parasitic oscillation” which is typically manifested as
bursts of high frequency oscillation (typically at 10MHz
or higher) superimposed on the audio signal.
Sometimes parasitic oscillation in Mosfets can be at
such a high frequency that it will not be seen on typical
20MHz oscillo
scopes; 100MHz or higher bandwidth
scopes are necessary to show it.
However, even though it may be invisible on a typical
oscilloscope, it is most important to stop it happening
because paradoxically, even though it is at such a high
frequency, it will cause the harmonic distortion to be much
higher than it otherwise would be and the amplifier will
sound unpleasant as a result. Anyhow, that’s why the
stoppers are included.
Capacitors C16-C19 are included to match the input
capacitance of the n-channel devices to that of the p-channel types. This improves the gain linearity at high audio
frequencies.
The 0.22Ω 5W resistors in series with the source of each
Mosfet are there to provide a degree of local negative feed-
back and to help improve the current
sharing between devices.
Trimpot VR1 is connected between
the collectors of Q9 and Q8 and is there
is provide a voltage offset between the
gates of the n-channel devices at the
top and the p-channel devices below.
This voltage offset becomes a forward
bias which turns on the Mosfets slightly in the absence of any audio signal.
This quiescent (ie, no signal) bias is
necessary to operate the Mosfets in
the more linear region of their transfer curve and thus reduces crossover
distortion.
Zener diodes ZD1 & ZD2 and diodes
D3 & D4 protect the gates of the Mosfets
from overdrive. The zeners and diodes clamp the drive voltage between
gate and source of each Mosfet to a
maximum of about +12.7V. Since the
Mosfets act as source-followers you
might wonder how the gate voltage
could go this high.
Normally, the peak current (at full
power into a 4Ω load) would be no
more than about 3-4A. Since the
transconduct
ance of these Mosfets
is about 1 Siemen or 1V/A, then the
gate-source voltage can be expected
to rise to no more than about 4V or
so under normal drive. So how could
the gate voltage ever rise much above
this figure?
The answer is that the gate drive
becomes excessive when the load of
the amplifier is short-circuited and it
is being driven hard. Under these conditions, the gate voltage to the Mosfets
could easily rise above 20 volts.
However, the zener diodes do not
provide short-circuit protection to
the amplifier. That is provided solely
by the fuses. The Mosfets are rugged
enough to withstand short circuits
until the fuses blow.
Negative feedback is applied from
the output of the amplifier, via R21, a
22kΩ resistor, to the base of Q5, part
of the first differential pair. The AC
gain is set by the ratio of the 22kΩ and
1kΩ resistors at the base of Q5 and this
gives a value of 23. The resulting input
sensitivity of the amplifier is 1.7V RMS
for 200 watts into 8Ω and 1.6V RMS
for 350 watts into 4Ω.
The low frequency response of the
amplifier is set by two time-constants.
The first is made up of the 1µF input
capacitor C1 and the 47kΩ input bias
resistor R3, giving a -3dB point of
3.3Hz. The second time-constant is
provided by the 1kΩ feedback resistor
Fig.6: this power amplifier is unlike most direct-coupled circuits in that it has
three differential stages to give it high open-loop gain before negative feedback
is applied.
August 1996 33
hot. Choke L1 is wound with 20.5
turns of 0.8mm enamelled copper
wire onto a 14mm plastic former.
Once it is wound, scrape the enamel
off the wire ends and then tin them
with solder before installing the
choke on the board.
When installing the fuse clips, take
note of their little lugs which should be
on the outside ends of the fuse.
Heatsink bracket
Fig.7: this diagram shows a suggested power supply for the amplifier. The
power transformer is rated at 500VA.
R19 and the 100µF capacitor C8, giving a -3dB point of 1.6Hz. Combined,
they result in a response which is only
-0.7dB at 10Hz.
At the high frequency end, the
main determinant of the response is
the double time-constant provided by
the input network consisting of R4,
R5, C2 & C3 which produce a rolloff
above 80kHz. Other factors affecting
the high frequency response are the
10pF capacitor shunting the feedback
resistor R21 and the output coupling
network consisting of R30, R31, L1 &
C10. The latter network is included to
ensure stability of the amplifier under
reactive load conditions.
Power supply
And now a few words about the
power supply.
Ideally, you need a supply which
can deliver over 300 watts if you are
using an 8Ω load and almost 600 watts
if you are using a 4Ω load. A good
compromise is to use a 500VA transformer and six 10,000µF capacitors,
as shown in Fig.7.
Note that the DC supply rails are
±70V, a total of 140V between rails.
This is a potentially lethal voltage so
be very careful when making measurements around the circuit!
Construction
As supplied in the kit from Altron
ics, the PC board has a solder mask and
screen printed component overlay, to
make assembly straightforward. The
component overlay diagram is shown
in Fig.8.
Start construction by fitting the PC
pins and the resistors, then install the
diodes, capacitors and small signal
transistors. Watch the orientation of
the electrolytic capacitors, diodes and
transistors. Don’t confuse the 1N914s
and 12V zener diodes.
Mount the 5W resistors about 3mm
off the PC board, just in case they get
The next task is to assemble the
MJE340 and MJE350 transistors onto
the heatsink bracket. A total of six
transistors need to be mounted. If
you hold the bracket so that it’s facing
you, three MJE350s (Q2, Q3, Q1) are
mounted on the left, then the two
MJE340s (Q6, Q8) and then another
MJE350 (Q9). It is most important not
to mix them up.
We should also make a note about
the brand of MJE340s and 350s. As
we have stated in the past, Motorola
devices are the best. Other brands will
work but they are nowhere near as
good, giving rise to less power output
and higher distortion.
Fig.9 shows the details of how each
MJE340 and MJE350 is mounted to the
heatsink bracket. You can use mica
washers and heatsink compound for
each transistor or use silicone impregnated thermal washers. Do not
overtighten the mounting screws.
When all six TO-220 transistors
are mounted on the bracket, it can
be installed on the PC board and the
transistor leads soldered.
An easier method is used to secure
the power Mosfets to their heatsink
RESISTOR COLOUR CODES
❏
No.
❏ 1
❏ 2
❏ 4
❏ 1
❏ 1
❏ 2
❏ 2
❏ 8
❏ 2
❏ 1
❏ 4
❏ 1
❏ 1
❏ 1
34 Silicon Chip
Value
470kΩ
47kΩ
22kΩ
4.7kΩ
3.3kΩ
1kΩ
470Ω
390Ω
270Ω
150Ω
100Ω
10Ω
4.7Ω
1Ω
4-Band Code (5%)
yellow violet yellow gold
yellow violet orange gold
red red orange gold
yellow violet red gold
orange orange red gold
brown black red gold
yellow violet brown gold
orange white brown gold
red violet brown gold
brown green brown gold
brown black brown gold
brown black black gold
yellow violet gold gold
brown black gold gold
5-Band Code (1%)
yellow violet black orange brown
yellow violet black red brown
red red black red brown
yellow violet black brown brown
orange orange black brown brown
brown black black brown brown
yellow violet black black brown
orange white black black brown
red violet black black brown
brown green black black brown
brown black black black brown
brown black black gold brown
yellow violet black silver brown
brown black black silver brown
PARTS LIST
1 PC board, code PEDK5180,
205 x 97mm
4 3AG fuse clips
2 5A 3AG fuses
1 large heatsink bracket
1 large single sided heatsink
1 small heatsink bracket
8 TO-3P mica insulating washers
6 TO-220 mica insulating washers
4 transistor mounting clips
7 PC pins
1 plastic bobbin
1 1.2m length of 0.8mm enamelled copper wire
1 200Ω horizontal trimpot (VR1)
Semiconductors
4 ECX10N20 n-channel Mosfets
(Q12,Q13,Q14,Q15)
4 ECX10P20 p-channel Mosfets
(Q10,Q11,Q16,Q17)
4 MJE350 PNP driver transistors
(Q1-Q3,Q9)
2 MJE340 NPN driver transistors
(Q6,Q8)
3 BC546 NPN transistors
(Q4,Q5,Q7)
4 1N914, 1N4148 signal diodes
(D1-D4)
2 12V 400mV zener diodes
(ZD1,ZD2)
Capacitors
2 100µF 160VW electrolytic
1 100µF 25VW electrolytic
1 1µF 63VW electrolytic
1 0.22µF metallised polyester
2 .047µF monolithic
1 .001µF greencap
1 470pF disc ceramic
1 330pF disc ceramic
1 220pF disc ceramic
4 22pF disc ceramic
1 10pF disc ceramic
Fig.8: the parts overlay for the PC board. Note that the 5W resistors should
be spaced 3mm off the board. Take care to ensure that all polarised parts are
correctly oriented.
bracket. Spring clips are used to clamp
adjacent transistors. The screw which
retains the spring clip also secures the
heatsink bracket to the PC board. A
cross-section diagram of the mounting
is shown in Fig.10.
All eight Mosfets are soldered to the
PC board first, making sure that there
is about 8mm of lead length above the
board. This allows them to be bent over
without placing too much strain on
the leads. When the eight Mosfets are
soldered in place, the heatsink bracket
and spring clips can be assembled
together. Do not forget to use a mica
washer and heatsink compound for
each device.
Place a spring clip over two Mosfets
Resistors (0.25W, 5%)
1 470kΩ
8 390Ω
2 47kΩ
2 270Ω
4 22kΩ
1 150Ω
1 4.7kΩ
4 100Ω
1 3.3kΩ
1 10Ω
2 1kΩ
1 4.7Ω 1W
2 470Ω
1 1Ω 1W
8 0.22Ω 5W wirewound
4 zero-ohm links
2 100Ω 5W (for biasing setup)
Miscellaneous
Screws, nuts, washers, solder,
heatsink compound.
August 1996 35
and then, using a 4mm screw from under the board, secure it to the heatsink
bracket. The screw for each clip should
be fully tightened; the beauty of these
spring clips is that you cannot apply
too much force to the Mosfets.
Make sure that all devices are
insulated from the heatsink bracket.
Check that all six TO-220 devices are
insulated from their heatsink bracket
as well.
Now check over all your assembly
work, making sure that the component
installed in each position agrees with
that on Fig.8.
Setting up and testing
You will need a power supply (see
Fig.7), a multimeter and a small screwdriver to set up the module.
Remove the two fuses and solder a
100Ω 5W wirewound resistor across
each fuseholder. Rotate trimpot RV1
fully anticlockwise. This setting will
Fig.9: here’s how the six TO-220 transistors are
mounted on the heatsink bracket. You can use mica
washers and heatsink compound for each transistor
or silicone impregnated thermal washers.
36 Silicon Chip
result in the minimum quiescent current through the output stage. Connect
the ±70V supply rails and ground to
the board. Don’t connect a signal or a
load at this stage.
Set your multimeter to DC volts
and connect it across one of the 100Ω
resistors on the fuse clips. Now switch
on. No smoke? Good! If all is not well,
switch off immediately!
Assuming no smoke, measure the
voltage across the 100Ω fuse clip
Fig.10: this diagram shows the mounting details
for the power Mosfets. Spring clips are used to
clamp adjacent transistors.
Kit Availability
resistor. It should be quite low, about
1V or so. Now rotate trimpot RV1 anticlockwise until the meter reads about
7V. This means that the output state
quiescent current is 70 milliamps.
Now measure the voltage across the
other 100Ω fuse clip resistor; it should
be about the same. Next, measure the
voltage across the speaker outputs.
The voltage can positive or negative
but should be less than 50mV.
Let the amplifier run in this condition for 10 minutes or so, to let the
bias stabilise. Re-measure the voltage
across the 100Ω resistors and adjust
trimpot RV1 if necessary.
The next job is to fit the amplifier
with a suitable heatsink and mount it
inside a case with a cooling fan and
power supply. You can then connect
a loudspeaker and signal source and
listen to your heart’s content.
Troubleshooting
If the 100Ω resistors smoked when
power was applied, then check the
following:
(1). Bias pot turned wrong way (should
be anticlockwise).
(2). Power Mosfets transposed (N types
with P types).
(3). Power supply wrongly connected.
(4). Short on underside of PC board.
(5). Output device(s) shorted to heat
sink(s).
(6). Shorted capacitor on power
supply (check greencaps and electro
lytics).
If the current is unstable (ie, jumps
all over the place), or the sound us
crackly or hissy, then the amplifier
is possibly unstable. Check the following:
(1). Wrong values of resistors in the
signal section (check them all).
(2). Ceramic capacitors are incorrect
value.
(3). Earth or ground connection missing.
SC
(4). Mosfet shorted to heatsink.
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August 1996 37
SATELLITE
WATCH
The spectacular failure of the first flight of the
Ariane 5 launcher on June 5th proved just how
vulnerable new launch technology can be. The
accident is the worst in European space history,
with a total estimated loss of US$500 million.
Despite the loss, industry experts
expect the long term effect on Ariane
space will be small. Flight V501 was
conducted by the European and French
space agencies and carried four Cluster
magnetospheric research satellites.
The successful launch of Palapa C2
on May 15 promises to breathe new life
into Indonesia’s satellite system. The C1
satellite, suffering some kind of power
problem, presently radiates vertically
polarised signals at a much lower level
than was anticipated. The C2 satellite,
observed in a test location of 124° E
longitude during June, is destined to
replace C1 as early as July at 113° E.
ASIASAT 2 – 100.5° E longitude: the
5-channel European bouquet of channels, marketed under the Deutsche
Welle banner, began operation in late
May. However, changes to the bitstream
made in June, allowed the first reception on domestic MPEG 2 equipment.
Presently, three of the five channels
are operational, providing broadcasts
from Germany, Spain and France. Yet to
come on line are MCM (France) and Rai
International (Italy). MPEG 2 decoders
should be available by the end of July.
Compiled by GARRY CRATT*
Meanwhile, several more analog
signals have become available on this
satellite at 1310MHz and 1425MHz IF.
Both signals are broadcast in Mandarin.
Releasing expansion plans, Asiasat
advise that their AS 3 satellite, carrying higher power transponders on both
C and K band, will increase coverage
by 16%. The satellite will be located at
the same location as Asiasat 1 (105.5°
E) which will be relocated to 122° E.
PALAPA C1 – 113° E longitude: although some improvement in signals
levels of vertically polarised transponders has been reported by enthusiasts in Australia and New Zealand,
levels are still below those originally
predicted. The replacement of this
satellite sometime in July should result
in greatly improved signal levels. It is
anticipated that parallel programming
will operate on the C1 and C2 satellites
for a few months after C2 location.
JCSAT 3 – 128° E longitude: after
months of testing, Japan’s newest DTH
operator “PerfecTV” commenced an
initial subscription drive in Japan.
Whilst designed only to be received
in Japan, the hardware cost for the
70-channel service is around US$700
and a three month free trial period will
run from June to September. PerfecTV
is a joint venture between Itochu Corp,
Mitsui & Co. Ltd, Nissho Iwai Corp and
Sumitomo Corporation.
Japan Satellite Systems has also
ordered a 4th satellite from Hughes
Space and Communications. JCSAT4
will be a HS601 satellite, similar to
JCSAT3 and will carry data, voice and
television signals to Japan and will be
equipped with multiple beams covering India, Australia and New Zealand.
The Eastern beam will cover Hawaii.
JCSAT1, which commenced operations in April 1989, will be retired in
August this year, after the discovery of
a fuel leak in March. The leak means
that the satellite will be retired two
years earlier than planned. As a temporary measure, JCSAT4 will be moved
to 150 E longitude, the same orbital location as JCSAT1, to ensure that there
is no service interruption. JCSAT1 is
presently used for DTH broadcasting,
news gathering services, VSAT, CATV
and vehicle tracking operations.
GORIZONT 41 – 130° E longitude: Raj
These three screen shots are from the new 5-channel European service carried by Asiasat 2.
38 Silicon Chip
*Garry Cratt is Managing Director of
Av-Comm Pty Ltd, suppliers of satellite TV
reception systems.
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TV continues to be the only transponder viewable in Australia from this satellite. A 1.8-metre dish is sufficient for
noise free reception anywhere along
the east coast. The IF is 1465MHz and
the polarisation is lefthand circular.
GORIZONT 42 - 142.5° E longitude: the
latest channel to commence operations
on this satellite is Indian broadcaster’s
“Global TV” adult channel, Channel
21. The nature of the programming and
the type (if any) of encryption remains
unknown. Typically, broadcasters will
operate for several months “free to air”
before offering subscriptions.
This brings the number of signals
that can be seen across Australia from
this satellite to three: EM TV New
Guinea, Asia Music/Zee Education
and Global TV. The IF for the global
service is 1375MHz. All services operate lefthand circular polarisation.
PANAMSAT PAS-2 - 169° E longitude:
July 1 saw the introduction of a new
analog service on this satellite. Located
at IF 965MHz, “The Value Channel”
operates in NTSC and will remain
unencrypted for the foreseeable future. The broadcaster offers domestic
viewers the opportunity to purchase
goods by telephone, using a credit
card. For domestic viewers on the east
coast of Australia, a 3-metre dish is
recommended.
As part of the Panamsat upgrade to
a DVB-compliant MPEG platform, a
number of broadcasters, previously
operating in the proprietary Scientific
Atlanta MPEG 1.5 standard (such as
CMT, ANB, CTN and CNBC) have been
relocated to allow simultaneous operations in both MPEG1.5 and MPEG2.
The 1.5 services were scheduled for
deletion on June 30, subject to completion of the rollout of replacement
D9223 IRDs. The new IFs for those
broadcasters are: CMT, ESPN-2, BBC
World, Bloomberg TV, 1249MHz hori
zontal and CNBC 1057MHz vertical.
INTELSAT 511 - 180° E longitude: this
satellite is scheduled to be changed to
Intelsat 701 (presently located at 174°
E), operating in geostationary orbit,
in late 1996. This will eliminate the
need for the autotracking equipment
now required for Intelsat 511. In recent
months, there has been an increase
in the number of broadcasters testing
MPEG circuits on this satellite. SC
August 1996 39
SERVICEMAN'S LOG
How many symptoms from one fault?
I believe it was Henry Ford who made the profound (?) statement that “history is bunk”. But
someone else, whose name escapes me, made
the rather more realistic statement that “he who
ignores history will be made to relive history”.
What has all that to do with servicing TV sets? Well, it turned out to
be singularly appropriate in regard to
the story I’m about to relate, though
I doubt whether either of the afore
mentioned philosophers was thinking
of anything so trivial (to them) as TV
servicing.
It all started with a Teac CT-M515S
colour TV set, a 51cm model about
three years old and featuring stereo
sound, Teletext, and remote control.
The owner’s complaint was straight
forward enough; it was completely
dead. And so it appeared to be at
switch-on – no sound, no picture and
no light on the screen.
Until I advanced the brightness
control, that is. Then the real symptom became obvious. There was no
vertical deflection, the set displaying
the classic thin bright line across the
centre of the screen.
A routine voltage check immediately produced a vital clue – there was
no 12V rail. The 12V rail is derived
from pin 4 of the horizontal output
transformer (T402) via a 0.68Ω fusible resistor (R423), diode D404, a
6.2Ω 3W resistor (R422), zener diode
ZD402, and a 1000µF filter capacitor
(C421).
The immediate cause of the supply
rail failure was ZD402, which had
broken down and taken out R422. But
that was not all. The vertical output
IC (IC401 - TDA2653B) had also been
destroyed. Which had come first and
destroyed which? There were no clues
on this but I regarded it as of secondary
Fig.1: part of the horizontal output circuit in the Teac CT-M515S colour TV
set. A 12V rail is derived from pin 4 of the horizontal output transformer,
via 0.68Ω fusible resistor R423, diode D404, 6.2Ω resistor R422, zener
diode ZD402, and 1000µF filter capacitor C421.
40 Silicon Chip
importance anyway. More to the point,
replacing those three components
was all that was needed to get the set
working again.
And it worked very well. I gave it a
thorough once-over, made some minor
setting-up adjustments, let it run for
a couple of days, and then returned it
to the customer. And that should have
been the end of the story.
Here we go again
It wasn’t, of course. A month went
by and the set was back in the shop.
Well, that was bad enough but the
really nasty part was that it was the
same components which had failed.
Which meant that I had treated only
the symptoms, not the cause. And I
had to find the cause.
I replaced all the damaged components again (it was becoming a costly
exercise) and the set came back to
life. But of course I couldn’t leave it
at that; I had to find what caused all
this destruction.
In general terms, I suspected an
over-voltage condition of some kind,
either high amplitude short term,
or lower amplitude continuously. I
couldn’t do much about checking for
the former but at least I could check
the latter.
So I made a complete voltage check,
looking for any values which were
even marginally high. This achieved
nothing directly; all values were virtually spot on.
But it did help indirectly, even
though I did the right thing for –
initially – the wrong reason. While
making these checks, I paid particular
attention to the high tension rail. This
rail is derived from pin 5 of the switchmode transformer (T901) via D904 and
normally sits at 113V, as measured at
test point TPB+.
However, between diode D904
and TPB+ there is a network of three
transistors: Q907 which is directly in
the HT rail line, Q906 which controls
Q907, and Q905 which controls Q906.
I didn’t recognise this network immediately. I assumed it was a voltage
regulator and, on this basis, wondered
whether a fault here could have been
responsible.
I was clutching at straws but decided to check all three transistors. And I
struck oil! Q907 was short circuit. But
gratifying though this was, it didn’t
altogether make sense. If it had ceased
to function as a voltage regulator, why
did the rail still measure 113V? Why
hadn’t it gone high?
I took another look at the network
and realised my mistake. It wasn’t a
voltage regulator at all. Instead, it was
a switching network, used to switch
the set on and off via the remote control system.
In greater detail, the switchmode
supply runs continuously while ever
the mains supply is on. The remote
control switches Q905 which in turn
switches Q906 and ultimately Q907 in
the HT rail to turn the set on and off. In
addition, the remote control switches
various signal paths.
It would be no problem to replace
the transistor but would this bring
me any closer to the real problem?
Well, it did. Deep down in the brain
cells, something stirred. Mr Ford’s
disparaged history was proving to be
anything but “bunk”. Rather, a whole
lot of historical bits were coming
together. So much so that I began deriding myself for not realising sooner
what might be wrong.
I went straight to C909, a 47µF 25VW
electrolytic on the base of switching
transistor Q904 in the power supply,
and reefed it out. I replaced it with a
high temperature, higher voltage type
and modified the mounting somewhat
to keep it as clear as possible from
heat sources.
Been there, done that
So what was the connection? This
switchmode power supply is virtually
identical to one produced by Siemens
many years ago – almost back to the
beginning of colour TV in Australia
– and which has been used by many
manufacturers since then. It was used
in some early HMV receivers (C211,
C221 series, etc) and more recently
in the Fujitsu-General FT-1411 and
FT-2011 receivers and the Sanyo
CTP6626, among others.
And it was memories of the Fujitsu-General FT-1411 which stirred first.
It all happened many years ago and,
Fig.2: the switchmode power supply in the Teac CT-M515S. The HT rail
comes off pin 5 and goes to switching resistor Q907 at top right. C909 is
at lower left.
what with my memory cells being
somewhat sluggish these days, I had
completely forgotten it. But as I recall
it now, the complaint was that it could
not be switched off properly via the
remote control.
And I use the customer’s term
“properly” because, while there was
no picture, there was still a raster on
the screen; ie, full line structure but
no video. It looked a simple enough
problem initially. The setup was almost identical with that of the Teac – a
transistor (Q606) in the HT rail (109V),
controlled in turn by Q605 and Q608,
the latter fed from the remote control
system. And Q606 had gone short
circuit. (As an aside, Q606 was a type
2SC2335, which is the same type as
Q906 used in the Teac).
Anyway, the problem was easily
fixed – a new 2SC2335 and the set was
back to normal. The trouble was, the
set bounced. I thought it was just bad
luck the first time but when it bounced
again I knew I was in strife.
I won’t bore the reader with all the
details as to how I finally cracked it
but, as I recall, it was a combination of
good luck and some physical evidence.
The physical evidence was signs of
corrosion around two electrolytics
in the power supply, C607 and C620,
both 100µF/25V. They were connected
in series to give 50µF and fed the base
of the switching transistor, Q604, in
August 1996 41
was all that was needed to put the sets
back in operation and minimise the
recurrence of the fault.
Which is pretty much where we
came in. And, no Mr Ford, history isn’t
bunk; it’s a very good teacher.
The money-hungry customer
the same manner as C909 in the Teac.
I can’t explain the reason for the
series arrangement. If it was to increase
the voltage rating, it wasn’t a very good
effort; there was no resistor network
to equalise the voltage distribution.
Anyway, I substituted a 47µF capacitor
with a higher voltage rating and a high
temperature rating and that finally
solved that problem.
Then there was the Sanyo CTP6626
which uses an 80P chassis (or more
correctly, there were several sets with
the 80P chassis). And, once again, this
uses what is virtually a Siemens type
switchmode power supply. In fact,
this story goes back even further and
would have been my first encounter
with this particular fault.
In this case, however, the story of
one fault is essentially the story of
them all. Apart from minor variations
42 Silicon Chip
(some sets were intermittent), they all
produced the same symptoms from the
same fault. It was real beaut at the time
because I quickly learned to handle the
situation. But it did little to prepare me
for the variations on the theme which
occurred in the Fujitsu-General and
the Teac several years later.
In essence, the problem presented
itself as a destroyed horizontal output
transistor (Q451), caused by a dramatic
rise in the main HT rail due, in turn,
to the failure of capacitor C314. C314
was a 47µF electrolytic capacitor in
the power supply and fed the base of
the switching transistor (Q304). And,
in some cases, Q304 would also be
destroyed.
Replacing the faulty transistors
and substituting a high temperature
electrolytic, mounted as far away as
possible from any sources of heat,
My next story, as fate would have it,
is also about a Sanyo TV set: a fairly
old set, a model CCC-3000, a 34cm
“Cosmo” portable, using an 80P chassis and, yes, the same Siemens type
power supply. But the story is just
about as far removed from the power
supply problems as it could be.
The customer was a European
gentleman with only a limited grasp
of English. But his grasp of money
matters suffered no such limitation;
he was as sharp as they come. So this
story is nearly as much about customer
relations, charges and the eternal problem of quotes, as it is about technical
problems. Inevitably, of course, the
two are interwoven.
The basic problem was simple; the
set had been dropped. Not particularly
hard apparently – there was no obvious external damage – but enough to
put it out of action.
Right from the start, and simply
on the basis that the set had been
dropped, the gentleman wanted me to
quote him to repair it. As a basic rule,
I don’t quote for repairs and certainly
not on the basis of such vague information. I will try to assess a particular
situation, based on the best available
evidence, but at best this is a guest
imation. There must inevitably be a
number of “ifs” and “buts” included
in such an assessment.
As a colleague once put it, “you
don’t really know what a job is going
to cost until it’s finished – and it’s a
bit late then to quote for it.” An exaggeration? Perhaps, but there is lot of
truth in that too.
Anyway, I explained that could not
quote him for the job and set out some
of the reasons. I told him I charged
so much an hour for labour, plus the
cost of any components which had to
be replaced.
The best I could do was switch the
set on and try to assess how much
damage had been done and, therefore
what kind of cost might be involved.
And, as I pointed out to him, I didn’t
even know whether the picture tube
was still working.
This didn’t seem to make much
Fig.3: the switchmode power supply in the Sanyo 80P chassis. It’s similarity
to the Teac supply is evident, both being based on an early Siemens circuit.
impression but I switched the set on
anyway. The result was more promising than I had expected. The sound
came up immediately and, as the tube
warmed up, there was some signs of
life on the screen – a bright horizontal
line. Well, this meant that the tube was
intact, the power supply was working,
and the horizontal output stage was
working. In fact, most of the vital
parts were working except the vertical
output stage.
On this basis, I told him that I
thought the most likely fault was a
cracked board. I couldn’t say how
serious this might be. It might be
possible to repair it or, if it was too
badly damaged, the only alternative
would be to replace it – assuming a
replacement was available and the
cost could be justified.
My most favourable assessment,
therefore, was that it would involve
at least two hours work. And that assumed that no components had to be
replaced, which I felt was a fair bet.
That still wasn’t good enough; he
insisted that I open the set, on the
spot, determine the exact nature of the
damage, and give him a firm quote for
a repair. I was equally insistent that
this was out of the question and that
the situation was not negotiable; take
it or leave it.
He hummed and hawed about this
but we finally reached a compromise.
I agreed to quote him for two hours
labour. If the job was going to cost more
than that I was to contact him and give
him the choice of either going ahead
with the job or aborting it, in which
case there would be no charge.
I wasn’t particularly happy with this
arrangement but felt fairly confident
that I could work within it. So it was on
to the bench and off with the back. My
immediate impression was that it had
obviously spent most of its life near the
ocean, because there was considerable
corrosion on the metal parts.
But I was looking for cracks. There
was nothing immediately obvious and
I removed the main board for a closer
inspection. The high risk areas would
be near the horizontal transformer and
where the board is supported by the
cabinet.
And this was where I found it;
from the transformer to the edge of
the board. It was a very fine crack,
about 10cm long, and not at all easy
to see. In fact, I suspected that at least
some of the copper tracks were still
be functioning, though obviously not
very reliably.
Anyway, it looked like a fairly
straightforward job, despite the fact
that a number of tracks were broken.
There was a fair amount of work involved in cleaning the board of dust
August 1996 43
Serviceman’s Log – continued
and grime, plus the original green varnish, until I was back to bright copper.
Then it was simply a matter of flowing
solder over the breaks.
Sometimes, if a crack is bad enough,
I fit a wire bridge but I didn’t consider
it necessary in this case. In fact, the
end result was very satisfactory, both
visually and mechanically.
via a 6.8kΩ resistor (R410) to pin 15,
which is marked as 12.7V. And this
voltage was spot on.
There wasn’t much left to suspect,
except the IC itself. Had I destroyed
it in some way while making tests? I
hoped not but the only way to prove
the point was to replace it. I pulled it
out and, because of possible doubts
The big test
Unfortunately, when I switched
it on, the result wasn’t satisfactory
at all; the vertical deflection stage
was still not working. My first reaction was to suspect that a supply
rail had been lost, perhaps because
of a crack I had missed.
I pulled out the circuit diagram
and began checking all the rails
which, at least at their starting
points, appeared to be correct. So
I began tracing them. And, since
I don’t like running a set for long
periods with a fault like this, I
would switch it on briefly, check
a voltage, then switch off while I
lined up another check point.
Then suddenly, when I switch
ed the set on, the white line had
vanished. And not because I’d
cured the fault but because the set
was now completely dead. This
was a really revolting develop
ment; instead of finding the fault
I had seemingly created another
one.
I went over the board again,
looking for any missed cracks, but
drew a blank. There was still the
full 110V on the main HT rail from
the power supply but no secondary
voltages from the horizontal output
transformer. And the CRO confirmed
that there was no horizontal activity
of any kind; nothing at the output
stage (Q451) and nothing at the driver
stage (Q450).
Further checks revealed that the
voltage on Q450’s collector was high.
Instead of the indicated 64.7V, it was
sitting at the full rail value of 110V. It
was obviously turned off and the CRO
confirmed that it was not being driven
from pin 3 of IC401, which contains
both the horizontal and vertical oscillators.
So why wasn’t IC401 working? This
IC takes its voltage from the 110V rail
44 Silicon Chip
why were there no other symptoms
due to the cracks?
Without backtracking and identifying every broken track, I can only
guess. However, it is possible that
there were other symptoms which
were masked by the vertical failure.
There may have been no video or no
colour, for example.
And why did IC401 then suffer a
further failure? This may have been
due to my test routine but I don’t
think so. Closer examination of the IC
revealed quite a lot of corrosion on
the pins, particularly where they
enter the plastic body. The pins
were firm enough mechanically
but it’s possible that some corro
sion had made its way inside the
body.
As a check, I refitted the original
IC in the socket and gave it a bit
of a bashing for good measure.
But it was completely dead. As I
say, these are questions for which
I have no answer.
Ungrateful customer
about my diagnosis, fitted a socket to
the board and plugged in a replacement IC.
And that fixed it. Not only did the
set come back to life when switched
on but the vertical scan had also been
restored and we had a full picture on
the screen.
Unanswered questions
All of which leaves a lot of unanswered questions. If the vertical failure
was due to a fault in IC401, rather
than the crack in the board, what had
caused it to fail?
Was the set running when it was
dropped and was there a voltage surge
when the copper tracks fractured? And
More to the point, in practical
terms, the job had now gone outside the terms of the cost agreement. As well as the two-hour
labour charge – which had been
exceeded but which I would carry
– the customer was now up for an
extra $20 for the IC.
Sticking to the agreement, I
rang him and advised him of the
situation. More aggro; he didn’t
want to go beyond the origi
nal
labour charge. I refused to budge.
I pointed out that I had kept my
part of the agreement and it was
up to him to keep his. And I added
the clincher – if he didn’t want to
pay the extra $20 I would put the old
IC back in the set and he could come
and collect it, no charge.
That did it. Knowing that I had the
set running on the bench but that I
could easily disable it was too much.
He agreed to pay the extra charge,
albeit reluctantly.
It was over a week before he turned
up and during that time the set never
faltered. But would you believe it,
when he came to collect it, he tried to
beat me down again.
I didn’t even argue with him; I made
it clear it was take it or leave it. He
took it – and I hope I don’t see it or
him again. Some customers are really
SC
not worth the trouble.
SILICON
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electronic design, and applications.
The sixth edition has been expanded
to include chapters on surface mount
technology, hardware & software
design, semicustom electronics &
data communications. 63 chapters,
in hard cover at $120.00.
Silicon Chip Bookshop
Radio Frequency
Transistors
Newnes Guide
to Satellite TV
Installation, Reception & Repair.
By Derek J. Stephenson. First
published 1991, reprinted 1994
(3rd edition).
This is a practical guide on the
installation and servicing of
satellite television equipment. The
coverage of the subject is extensive, without excessive theory or
mathematics. 371 pages, in hard
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Guide to TV & Video
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By Eugene Trundle. First publish-
ed 1988. Second edition 1996.
Eugene Trundle has written for
many years in Television magazine
and his latest book is right up date
on TV and video technology. 382
pages, in paperback, at $39.95.
Servicing Personal
Computers
By Michael Tooley. First published 1985. 4th edition 1994.
Computers are prone to failure
from a number of common causes
& some that are not so common.
This book sets out the principles
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format and R-DAT. If you want to
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The Art of Linear
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This is a practical handbook from
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336 pages, in paperback at $49.95.
Components, Circuits & Applica
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1990.
Previously a neglected field, power
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Digital Audio & Compact
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Electronics Engineer’s
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This book strips away the mysteries of RF circuit design. Written
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Surface Mount Technology
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This book will provide informative
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This book is for anyone involved
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Title
Newnes Guide to Satellite TV
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The Art Of Linear Electronics
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Electronic Engineer's Reference Book
Radio Frequency Transistors
Surface Mount Technology
Audio Electronics
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Are your TV signals weak or
noisy? This masthead amplifier
could mean the difference
between a lousy picture and
good reception.
Portable
masthead
amplifier
for TV & FM
By BRANCO JUSTIC
T
HIS MASTHEAD AMPLIFIER
was originally designed for use
with caravans and recreational
vehicles. It’s portable, comes with its
own inbuilt telescopic (rabbit ears)
antennae and runs off a power supply
ranging from 7-20V DC or 6-15V AC.
This means that you can either power the unit from a 12V car battery or
from the mains via a suitable plugpack
supply.
The “rabbit ears” telescopic antennae feed directly into the amplifier
circuit. This circuit typically provides
from 16-20dB of gain at frequencies
up to 1GHz, which should be plenty
54 Silicon Chip
for beefing up an otherwise marginal
signal to a portable TV set.
If you’re fed up with constantly adjusting the antenna on your portable
TV set or if the reception varies when
you change channels, this “active”
antenna system is the way to go. It not
only amplifies the incoming signal but,
just as importantly, provides correct
impedance matching between the
antenna and your TV set.
Of course, there’s nothing to stop
you from using this design in fixed
installations or as a distribution amplifier. All you have to do is ditch the
rabbit ears antennae and feed a signal
in directly from a fixed antenna or a
distribution cable. The unit is easy to
install and is suitable for amplifying
both VHF and UHF signals, as well as
FM signals.
As with most masthead amplifiers,
the DC supply rails are delivered via
the downlead; ie, the TV signal and
the supply rails share the same cable. This means that you don’t have
to run separate supply leads up the
mast, which greatly simplifies the
installation.
Generally, the best approach is to
mount the amplifier as close to the
antenna terminals as possible. That’s
Fig.1: the circuit is based on a MAR6 broadband RF amplifier (IC1) which
provides around 20dB of gain. D1 and D2 protect the input of IC1 by clipping
any high voltage transients, while REG1 provides a 5V supply rail. This supply
rail is fed via the signal cable to the output terminal of IC1 and is isolated from
the TV set using C3.
Fig.2: install the parts on the two PC boards as shown here. The MAR6
(IC1) is mounted from the copper side of the board (see photo).
really just another way of saying that it
should go on the mast. This is done to
avoid signal degradation due to cable
losses. Quite often, a good signal is
available at the antenna terminals but
cable losses can result in a severely
degraded signal by the time it reaches
the TV set.
The basic idea is to amplify the good
signal that’s coming from the antenna,
rather than a noisy signal at the TV
set itself.
Well, that’s what the theory says.
In practice, you can sometimes get a
good result by placing the masthead
amplifier at the TV if you don’t want
to go to the trouble of mounting it on
the mast. This only applies to borderline situations, where the signal is just
too weak for the AGC (automatic gain
control) circuit to limit the front-end
gain of the receiver.
In this situation, you get a “snowy”
picture because the front-end operates
at high gain which results in a poor
signal-to-noise ratio. By amplifying the
signal before it is fed into the receiver’s
front end, the AGC circuit limits the
gain and this drastically cuts the noise
to give a clear picture.
Distribution amplifier
Another area where this circuit
should prove popular is as a distribution amplifier. Quite often, a signal
that’s adequate for one TV set will no
longer be adequate when fed through
a splitter for distribution to several
outlets. That’s because the splitter itself introduces signal losses, typically
around 3.5dB or more.
The amplifier board (left) is installed
inside a length of 100 x 43mm OD
conduit. Above is a close-up view of
the MAR6 IC, which is mounted on
the copper side of the board.
August 1996 55
The power supply board is installed inside a small plastic utility case, as shown
here. Take care to ensure that the 7805 regulator is oriented correctly and check
that the completed unit delivers +5V to the centre conductor of the lead that
runs to the amplifier board.
The answer is to amplify the signal
before feeding it to the splitter. Doing
this will ensure a sufficient level at
each outlet for a noise-free picture,
despite losses in the splitter circuit
and the distribution cable.
Circuit details
Fig.1 shows the circuit details. It’s
based on a MAR6 monolithic broad
band amplifier (IC1) made by Mini-Circuits (USA). This device has a rated
bandwidth from DC to 2GHz, 20dB of
gain at 100MHz and a low noise figure
of around 2.8dB. This noise figure is
far superior to the noise figure for the
OM350 mono
lithic amplifier used
in many older masthead amplifier
designs.
Apart from the MAR6, there’s just
Fig.4: a masthead amplifier is useful for boosting the signal
before it is fed to a splitter for distribution to multiple TVs.
Fig.3 a balun is necessary if you
intend using the twin telescopic
antenna. It is wound using lightduty single core wire.
a 7805 3-terminal regula
tor, three
diodes and a few minor components.
All the required gain is provided by
the MAR6, so there’s no need to make
things complicated. Let’s take a closer
look at how it works.
The signal from the antenna is coupled to the input of IC1 via capacitors
C1 and C2 which provide DC isolation.
Diodes D1 and D2 are there to protect
IC1 from excessive input voltages,
as could be induced by nearby RF
transmitters, lightning strikes or static
build-up.
Note that BAW62 diodes are specified here, as these are a high-speed
switching type with very low capacitance. As a result, they provide good
protection for IC1 with very little
signal loss. In operation, they clip any
high voltage spikes to ±0.6V.
The amplified signal appears at the
output of IC1 and is coupled directly
to the centre conductor of the coaxial
cable downlead. It is then subsequently fed to the antenna terminal of the
TV set via C3.
Power supply
Fig.5: here’s how to include a VCR in a distribution system.
The combiner is just a 2-way splitter wired back-to-front.
56 Silicon Chip
Power for the circuit is derived from
an external AC or DC plugpack supply.
D3 either rectifies the AC supply or,
in the case of a DC supply, provides
Do You Need A Masthead Amplifier?
“Will a masthead amplifier solve
my TV reception problems?”
That’s a question that’s often
asked and the answer is “it depends”.
A masthead amplifier is not a universal panacea for crook TV pictures
and there are many situations where
it will offer little or no improvement.
It will not eliminate most ghosting
problems, for example, as the
ghosts just get amplified along with
everything else.
Nor can a masthead amplifier
clean up interference problems or
give you a good picture if there is
little or no signal in the first place.
reverse polarity protection. The resulting DC rail is then filtered by C5 and
drives 3-terminal regulator REG1. The
5V output from REG1 is then filtered
and applied to the output terminal of
IC1 via R1, L1 and the centre conductor of the downlead.
Inductor L1 presents a high impedance at signal frequencies and thus
ensures that IC1’s output is not loaded
by the supply rail. It also serves to keep
signal frequencies out of the regulator
output circuitry.
Construction
The assembly of the masthead
amplifier is straightforward, with
all the parts mounted on two small
PC boards. The MAR6 RF amplifier
and its associated parts go on the
smallest board and the completed
assembly installed inside a length of
100 x 43mm OD plastic conduit. This
That said, there are many situations where a masthead amplifier
can dramatically improve picture
quality, particularly in fringe areas.
Basically, you should use a masthead amplifier under the following
circumstances:
(1) You live in a fringe area and
one or more channels is noisy;
(2) Reception is poor due to losses
in the downlead;
(3) The signal strength is inadequate because of splitter and cable
losses in a distribution system;
(4) The antenna system is only
very modest.
is fitted with end caps for weather
proofing – an important consideration
if the unit is to be mounted outdoors
on an antenna mast.
The power supply parts are accommodated on the second board. This
board fits inside a small plastic utility
case which would normally be hidden
somewhere behind the TV set.
Fig.2 shows the parts layout on the
two PC boards. Begin by installing the
parts on the amplifier board, taking
care to ensure that diodes D1 and D2
are oriented in opposite directions.
The two capacitors are non-polarised and can be installed either way
around.
The MAR6 amplifier IC is a surface
mount device and is installed from
the copper side of the PC board. The
accompanying photographs show
how this is done. Make sure that it
is correctly oriented. Its type number
Where To Buy The Parts
Parts for this masthead amplifier design are available from Oatley Electronics,
5 Lansdowne Parade, Oatley, NSW 2223. Phone (02) 579 4985 or fax (02) 570
7910. Prices are as follows:
Basic kit (incl. PC boards, MAR6 IC, all on-board parts & balun core) ....$15.00
Twin telescopic antenna .............................................................................$5.00
Plastic case for power supply .....................................................................$2.50
Plugpack supply .......................................................................................$10.00
RG59 coaxial cable..............................................................................90c/metre
Payment may be made by cheque or credit card. Please add $5 for packaging
and postage.
Note: copyright of the PC board artworks associated with this design is retained
by Oatley Electronics.
PARTS LIST
1 amplifier PC board (Oatley
Electronics)
1 power supply PC board
(Oatley Electronics)
1 twin telescopic antenna
(optional)
1 balun core
1 100mm length of 43mm O.D.
plastic conduit
2 43mm I.D. end caps
1 plastic case, 84 x 54 x 30mm
4 plastic cable ties
1 15µH inductor (L1)
1 68Ω resistor (0.25W)
Semiconductors
1 MAR6 wideband RF amplifier
IC (IC1)
1 7805 3-terminal regulator
(REG1)
2 BAW62 fast switching silicon
diodes (D1,D2)
1 1N4004 silicon diode (D3)
Capacitors
1 100µF 25VW PC electrolytic
(C5)
1 .0033µF ceramic (C4)
3 .001µF ceramic (C1,C2,C3)
Miscellaneous
Light-duty single core wire (to
wind balun), clamps, silicone
sealant, coaxial cable.
should be visible from the component
(top) side of the PC board, while a
small white triangle or dot indicates
the input pin.
If you are using 75-ohm coaxial
downlead from the antenna, this can
be soldered directly to the PC board as
shown in Fig.2. Alternatively, the rabbit ears antenna comes with 300-ohm
ribbon cable and so a balun is necessary to match this (and other standard
antennas which don’t already have a
balun) to the 75-ohm input impedance
of the amplifier.
Fig.3 shows the winding details of
the balun. It is wound using light duty
single core wire. The amplifier side
consists of a single turn through the
core, while the antenna side consists
of two turns wound from the opposite
end of the core.
On the prototype, the rabbit ears
antenna was mounted on one of the
end caps (see photo) and secured using machine screws and nuts. Once
August 1996 57
can now be installed in its case and
the external connections made. You
will need to drill holes in one end of
the case two accept the two coaxial
cables and the power supply leads. As
before, attach cable ties to the various
leads just inside the case so that they
cannot be pulled out.
Although not shown on the prototype, we recommend that the power
supply leads be run to a suitable jack
socket mounted on the end of the case.
That way, the plugpack supply can
be easily disconnected and used in
another application if required.
Installation
The plastic conduit case makes a
neat weatherproof assembly which
is easily attached to a mast using a
large hose-clamp.
the connections have been made, the
completed amplifier board is pushed
into its plastic conduit housing. The
output lead emerges through a hole
drilled in the bottom end cap.
For a fixed installation, the 75-ohm
antenna lead is fed in through a second hole in the bottom end cap. It’s a
good idea to fit a couple of cable ties
to the cables just inside the end caps
to provide strain relief for the soldered
connections. The two entry holes can
later be sealed with silicone sealant
after the assembly has been completed
and tested.
58 Silicon Chip
The power supply board can now
be assembled and tested. Take care to
ensure that D3 (1N4004), C5 and REG1
are all correctly oriented. Inductor L1
(15µH) looks like a resistor. It has a
light green body and carries brown,
green, black and silver colour bands.
Once the power supply board has
been completed, temporarily apply
power and check that the output side
of L1 is at +5V with respect to ground.
If there’s a problem here, switch off
immediately and carefully check the
circuit around REG1 and D3.
Assuming that all is well, the board
The way in which the unit is used as
an active antenna for portable TV sets
is obvious – just unplug the existing
antenna and plug this unit in instead.
Don’t forget to apply power to the
amplifier though.
For use with an outdoor antenna,
the amplifier unit should go up on
the mast as mentioned previously.
This arrangement will provide the
best signal-to-noise ratio although a
short length of high-quality coaxial
cable between the antenna terminals
and the masthead amplifier shouldn’t
make too much difference.
If used as a distribution amplifier,
the unit can be mount
ed indoors,
provided that input signal from the
antenna is noise-free in the first place.
The output of the amplifier is
connected to the splitter input and
the splitter outputs then run to the
various TV receivers. Fig.4 shows the
basic idea.
Finally, if strong signals on one or
more channels cause receiver overload (as indicated by an interference
pattern), try fitting a tuned attenuator
for the offending channel(s). This
should be fitted right at the antenna
terminals (ie, before the masthead
amplifier).
A 1/4-wave stub makes a very
effective tuned attenuator. This is
simply a length of coaxial cable cut
to exactly a 1/4-wavelength of the offending channel. If the stub attenuates
the signal too severely, try making
it slightly shorter until you get the
desired result.
Another approach is to initially cut
the stub slightly shorter than 1/4-wavelength and then tune it towards resonance using a trimmer capacitor across
the far end. Just keep on experimenting
SC
until you get it right.
SILICON
CHIP
If you are seeing a blank page here, it is
more than likely that it contained advertising
which is now out of date and the advertiser
has requested that the page be removed to
prevent misunderstandings.
Rod Irving Electronics Pty Ltd
SILICON
CHIP
If you are seeing a blank page here, it is
more than likely that it contained advertising
which is now out of date and the advertiser
has requested that the page be removed to
prevent misunderstandings.
Please feel free to visit the advertiser’s website:
Rod Irving Electronics Pty Ltd
SILICON
CHIP
If you are seeing a blank page here, it is
more than likely that it contained advertising
which is now out of date and the advertiser
has requested that the page be removed to
prevent misunderstandings.
Rod Irving Electronics Pty Ltd
SILICON
CHIP
If you are seeing a blank page here, it is
more than likely that it contained advertising
which is now out of date and the advertiser
has requested that the page be removed to
prevent misunderstandings.
Please feel free to visit the advertiser’s website:
Rod Irving Electronics Pty Ltd
SILICON
CHIP
If you are seeing a blank page here, it is
more than likely that it contained advertising
which is now out of date and the advertiser
has requested that the page be removed to
prevent misunderstandings.
Rod Irving Electronics Pty Ltd
The oscilloscope is a wonderful measurement
tool but if it is not used carefully it can give
highly misleading results. You can achieve the
full potential of your scope but only if you know
what you are doing. This article gives some good
tips on oscilloscope use.
By BRYAN MAHER
Say you have invested hard cash in
a good quality oscilloscope. It looks
a beautiful instrument and the specs
guarantee it to be accurate within
2%. Wow! And its bandwidth is wide
enough to make your friends drool.
But a scope is only a tool, no matter
how glossy the literature. If you don’t
use it properly you will be disappointed with the results.
Let’s start with a simple DC measurement, using the circuit shown in
Fig.1(a). If we read the DC voltage at
point D, a digital voltmeter (DVM)
gives a reading of +4.9V. If we then
connect the oscilloscope via a 1x
shielded probe, the deflection on the
screen is likely to indicate only about
+4.17V. Which is correct? Clearly that
64 Silicon Chip
scope probe is loading the source of
this measurement, pulling the voltage
down!
“Source” here means any part of
circuit at which we make a measurement. In this case it is point D in Fig.1.
And “source resistance” or “output
resistance”, denoted by Rs, means the
ratio of the change in voltage at that
point (caused by attaching the probe)
divided by the minute current drawn
by the probe.
This is denoted by the expression:
Rs = (∆v/∆i) Ω.
Because it is a voltage/current ratio, we call it resistance (ohms), even
though it is a calculated quantity.
Only rarely is Rs a single physical
component. Nevertheless Rs does have
the ability to upset the workings of a
circuit. “Delta” simply means a small
change in any quantity.
Equivalent circuit
The equivalent circuit, illustrated
in Fig.1(b), reveals how this loading
effect occurs. The input resistance of
the direct 1x probe connection is just
the 1MΩ resistor within the scope,
which we have called R1 in Fig.1(a).
R1 and Rs actually form a voltage
divider, so the scope sees only the
voltage at D, which is the true voltage
of the source reduced by the fraction
(R1/(R1 + Rs)).
Typically, a digital multimeter has
an input resistance of 10MΩ so using
it has a less deleterious effect on the
voltage. This is why the DMM reading
is higher, at +4.9V.
You can calculate the value of the
source resistance Rs in this case from
these measurements and the definition
given above. It works out to be about
200kΩ which is reasonable for this
particular op amp circuit.
Let’s define V as the unloaded
output voltage of the source; ie, the
potential at point D when neither the
scope probe nor the DMM is connected
to it. Using the voltage divider equa
tion, the voltage Vpat D when only the
1x probe and scope is hooked on is:
Vp = V(R1/R1 + Rs)
= V(1MΩ/1.2MΩ)
= V/1.2
The scope reads Vp as +4.17V, so
the unloaded output vol
tage at the
point D is:
V = (1.2)(4.17) = 5V
The relatively low resistance of the
scope input was the cause of the loading effect. It loaded the source and so
caused the oscilloscope to read +4.17V
instead of the true +5V.
Measurement rule-of-thumb
The cure for this loading effect is
now obvious. The test instrument
should have an input resistance much
greater (preferably 100 times greater)
than the output impedance of the
source to be measured. A 100 times
factor would limit loading errors to
about 1%. But practical aspects like
price, availability and frequency
response will limit our selection of
scope probes.
A common favourite, the 10x probe,
as illustrated in Fig.2, is an excellent
choice in most cases. This type of
probe contains a 9MΩ resistor called
Rp. Therefore the total probe connection resistance, Rin, is equal to Rp in
series with the scope input resistance,
R1. That is:
Rin
= (Rp + R1)
= (9MΩ + 1MΩ) = 10MΩ
If we substitute this 10x probe in
the measurement shown in Fig.1, the
oscilloscope would display a deflection of +4.9V, the same as the DMM
reading, a satisfying result.
Fig.1: this dual phase amplifier (a) has a 5V output at point D where the
source resistance is 200kΩ. But clipping the 1MΩ probe onto this point pulls
the voltage down to 4.17V. The equivalent circuit (b) shows that the source
resistance Rs forms an unwanted voltage divider with R1, the input resistance
of the 1x probe and the scope. This reduces the voltage seen by the scope.
High voltage measurements
Fig.2 shows a second important
use of the 10x probe. Here the source
resistance is quite low (due to negative
feedback) at the collector of transistor
Q1 so loading is not a worry but the
high voltages are! In this case we can
use the fact that Rp (in the probe head)
and R1 (in the oscilloscope) form a
deliberate voltage divider. Any voltage
which we apply to the probe tip will
be reduced at the scope input terminal.
The reduction fraction is:
Vsc = R1/(Rp + R1)
= 1MΩ/(9MΩ + 1MΩ)
= 1/10.
That’s why this probe is known as a
10x, because it produces a 10:1 voltage
attenuation.
In the circuit of Fig.2, the high volt-
Fig.2: the 9MΩ probe resistor Rp and the 1MΩ scope input resistor R1, form a
deliberate voltage divider. This reduces the voltage at the oscilloscope input
terminal to one tenth of that at the probe tip.
age of the supply (+450V) rules out
use of the 1x probe and forbids direct
connection to the scope’s input. But
the 10x probe is suitable, provided it
has a voltage rating above 450V. This
probe will reduce all waveform voltages to one tenth and the DC voltage
at the scope input will be no more
than +45V.
By dividing down the signal, the
10x probe effectively multiplies the
V/div calibration on the attenuator
switch by a factor of 10. So a 5V/div
setting now means 50V/div and eight
vertical divisions on the screen will
correspond to a 400V range. Hence
this 360V signal fits within the graticule limits.
Many top line scopes can sense
when the 10x probe is connected to the
modified BNC input terminals. Then
internal logic circuits multiply the
August 1996 65
Fig.3: source (a) has output resistance Rs equal to 50Ω at point D.
The high frequency equivalent circuit (b) shows that Cp forms an
unwanted voltage divider with Rs. Cx represents the combined
stray capacitance of the coaxial cable and the scope input.
10x probe) we must be aware that the
probe tip still carries a lethal 360V! For
safety we must keep the amplifier 0V
line connected to the scope frame and
to mains earth. And we never unplug
the probe from the scope while the
probe tip is still hooked onto a high
voltage point.
All probes which contain only resistors and capacitors are called passive and oscilloscope manufacturers
market a range of higher resistance
units. A few of these are listed in Table 1 but not all probes on the market
have voltage ratings as high as those
shown here.
Direct 1x scope probes have only a
small series resistance so they cause
little attenuation of the signal being
measured. They are useful for the
display of very small voltages of low
frequency signals, when measured at
low impedance points, such as the
outputs of op amps.
Some less common sources, like
biological assay electrodes, have an
extremely high output resistance.
To display signals from these, active
probes are required. Typically, these
employ IGFETs and other active circuitry to provide an input impedance
of 10GΩ and zero input capacitance.
Oscilloscope bandwidth
Fig.4: the amplitude response of an oscilloscope falls at high frequencies. At
full rated bandwidth, the response is -3dB or 30% lower than it is at low
frequencies.
on-screen readout by 10, to correctly
display the voltage value at the probe
tip. This facility is not provided in
cheaper scopes and nor does it work
when a scope is used with a probe of
a different brand.
Safety precaution
Though the oscilloscope is safely
working on reduced input voltages
(because of the attenuation by the
Fig.5: With AC (capacitive) coupling,
the signal passes through a high pass
filter. This will reduce the amplitude
of low frequency signals and distort
low frequency pulse waveforms.
Table 1
Probe
Attenuation
1x
10x
100x
1000x
R(in)
1M
10M
10M
100M
66 Silicon Chip
Maximum DC
Voltage
350V
600V
1.5kV
20kV
Derated Above
Derated to
1MHz
200kHz
100kHz
30V <at> 20MHz
300V <at> 20MHz
2kV <at> 20MHz
Another scope parameter which
new users often have difficulty coming to terms with is bandwidth. This
could be easily measured if you had a
synthesised RF signal generator with
an output of 5V over a frequency range
from 100kHz to 250MHz and an output
impedance of just 50Ω.
You might think that such a wide
band source could easily demonstrate
a scope’s bandwidth. Would you just
connect the 10x probe to the generator
and then sweep over the frequency
range? Fig.3 illustrates the setup, with
the probe’s internal resistance and
capacitance shown.
However, you might be disappointed to find that, when the generator
was set to the advertised bandwidth
frequency of your high performance
scope, say 250MHz, the vertical deflection is only half what it should
be. So what does scope bandwidth
mean?
The bandwidth of any oscilloscope
is that high frequency at which the
response has fallen to 70.7% (-3dB),
compared to the reference frequency
value, as illustrated in Fig.4. This
Table 2
Taken from a Tektronix TDS360 digital oscilloscope, this screen printout
shows the effects of incorrect adjustment of 10x probes on the scope’s internal
1kHz compensation signal. Channel 1, the upper trace, shows too much probe
capacitance (over-compensation) while the channel 2, lower trace, shows
insufficient capacitance (under-compensation). The correct probe compensation
adjustment would show a square wave with “square” corners.
This scope printout shows the effects of DC and AC coupling on a pulse
waveform with uneven duty cycle. Channel 1, top trace, is DC coupled and it
can be seen that the voltage swings equally above and below the zero reference
line (solid horizontal cursor). Channel 2, lower trace, is AC coupled and the
waveform has floated down with respect to the zero reference line (dotted
horizontal cursor).
shows that the response of any oscilloscope is down by 30% at its advertised
full bandwidth!
Furthermore, Fig.4 shows that the
manufacturer’s guarantee of an amplitude error of less than 2% only applies
for signal frequencies less than one
quarter of the rated bandwidth.
Frequency
Capacitive
Resistance
1MHz
10MHz
50MHz
100MHz
250MHz
300MHz
400MHz
13.3k
1.3k
265
132
53
44
33
Therefore, to make amplitude
measurements with less than 2%
error, we need a scope with a quoted
bandwidth four or five times higher
than the signal frequency. For example, accurate amplitude display of a
50MHz sinewave requires a 250MHz
oscilloscope.
This is only part of the bandwidth
story. As we noted above, testing an oscilloscope with a wideband generator
could show an error of more than 50%
at the advertised scope bandwidth.
How could it get worse?
In most cases the advertised -3dB
bandwidth of a scope applies only
when signals are coupled directly into
the instrument front terminal and not
via a probe, because probes also have
frequency limitations.
This is demonstrated by Fig.3(b),
which is the high frequency equivalent
of the circuit shown in Fig.3(a). As
before, the resistance presented by the
probe and scope connection is:
Rin = (Rp + R1) = 10MΩ
where Rp is the resistance inside the
probe and R1 is the input resistance
of the oscilloscope.
In the equivalent circuit of Fig.3(b)
we can ignore the 10MΩ input resistance Rin because it is so much higher
than the 50Ω source resistance Rs. But
we cannot discount the probe’s input
capacitance Cp which is equal to 12pF.
The capacitive reactance of Cp is:
Xc = 1/(2πfCp).
This forms an unwanted voltage
divider with the source resistance Rs.
At high frequencies the resulting low
value of Xc drastically reduces the
signal amplitude before it enters the
scope. Table 2 demonstrates the severity of this effect, with the reactance of
12pF at specific frequencies.
From Table 2, we observe that at
250MHz the probe’s capacitive reactance has fallen to 53Ω. Now we will
August 1996 67
Fig.6: since a PWM signal has a varying duty cycle and therefore an effectively
varying positive and negative DC offset, AC coupling will cause the waveform to
waver above and below the 0V reference line.
see the reason why the amplitude
displayed on the screen fell to 50%.
Firstly, looking at Fig.4(b), we see
that at 250MHz the voltage divider
effect of the 53Ω Xc with the 50Ω
source resistance Rs reduces the
signal voltage at D down to 70% of
the unloaded source voltage (it’s a
vector calculation, because of the
capacitor).
Secondly, as Fig.4 shows, the
displayed amplitude will be further
reduced to 70% of the voltage at the
scope input, because the signal frequency is now equal to the 250MHz
bandwidth of the scope. So the amplitude you would see on the screen
will be reduced to (70% x 70%) =
50% of the unloaded source voltage.
That explains why a high frequency
measurement with a 10x probe can
have such large errors.
Table 3
Attenuation
R(in)
C(in)
1x
10x
10x
100x
10x
10x
1M passive
10M passive
10M passive
10M passive
100k active
500 divider
55pF
12pF
8pF
2.7pF
0.4pF
0.15pF
68 Silicon Chip
Only in a few cases will a manufacturer guarantee that the advertised
bandwidth applies at a specified probe
tip. Examples include the Tektronix
400MHz oscilloscope model 2465B
but only when used with their 1MΩ
passive 10x probe model P6137.
Table 3 shows the input capacitance
and bandwidth of typical probes.
Frequency pulling
Often, the application of a passive
scope probe to some points of a circuit
can have drastic effects, particularly
in the case of crystal and other oscillators.
These require critical positive feedback gain and phase, set by specific
small capacitor values, to maintain oscillation at the required frequency. But
hooking a passive probe onto a high
impedance point of these circuits can
add 12pF of capacitance, upsetting the
feedback. This action
can either reduce the
Bandwidth
operating frequency
or may stop oscilla15MHz
tion altogether.
100MHz
How do we avoid
500MHz
this? Many systems,
250MHz
including some TV
4GHz
rec eivers, contain
buffered test points,
9GHz
where sensitive circuits are accessed
either via an inbuilt resistor or a low
impedance source follower.
Alternatively, a simple expedient is
to attach a small resistor, about 10kΩ,
to the probe and use the other end of
that resistor as the probe point. The results may be inaccurate but at least you
can monitor the waveforms. Another
alternative is to use a high impedance
active probe, such as listed in Table 3.
For frequencies above 500MHz,
wideband active FET probes are
available with a high input impedance
and they require a separate supply.
Examples include the Tektronix type
P6204 which has a 1GHz bandwidth
and the type P6217 which operates
to 4GHz. Active probes accept small
input voltages, typically below 10V.
For really wide bandwidth scopes,
between 2GHz and 10GHz, low impedance divider probes are available, with
input resistances of 50Ω, 500Ω or 5kΩ.
They plug into the 50Ω input terminals
on very high frequency oscilloscopes.
Probe risetimes
Another area where a new oscilloscope can disappoint is when displaying square waves which are supposed
to have fast rise and fall times. Fig.3(a)
shows the connection as before and
now we will explain why the probe
capacitor Cp is there at all, in view of
the trouble it causes when displaying
very high frequencies.
The reason why Cp is inside the
and undershoot. Naturally this Cp
adjustment also has a big effect on the
displayed bandwidth so if you don’t
adjust it correctly, it is yet another
source of measurement error.
AC coupling
This scope printout shows the effects of DC and AC coupling on a pulse
waveform with varying pulse width (ie, pulse width modulation). Channel 1,
top trace, is DC coupled while Channel 2, the lower trace, is AC coupled. The
varying pulse width effectively becomes a varying DC offset which is reflected
as a wavy modulation on the waveform, an erroneous display. This is the same
effect as depicted in Fig.6.
probe becomes clear when you look at
pulse risetimes. The probe’s shielded
cable and the oscilloscope’s input
stage add up to a considerable capacitance to ground, probably between
35pF and 100pF.
This we denote as Cx in Fig.3(a).
If Cp did not exist in the probe head,
then the probe resistor Rp, together
with this stray capacitance Cx, would
form a severe low pass filter. The
effect would be a reduction in amplitude and a phase change in sinewave
signals and a drastic slowing of the
risetime of pulses as displayed on
the screen.
Therefore the capacitor Cp has been
deliberately included in the probe to
correct these errors. But Cp must be
correctly adjusted until the two time
constants, RpCp and R1Cx, are equal.
To facilitate this adjustment, most
oscilloscopes provide a fast-risetime
1kHz square wave calibrating signal
from a terminal (usually) on the front
panel. You just hook the probe onto
this CAL terminal and adjust the probe
capacitor Cp until the scope displays
a true square wave.
If Cp is set too low, the square wave
will be rounded off while if Cp is too
high, the square wave will overshoot
So far we have talked about large
DC voltages and high frequencies but
if you have a circuit with high DC
voltages and small signals, you need
to switch the scope’s input to AC cou
pling. This enables you to use high
input sensitivity while blocking out
a large quiescent DC voltage. As Fig.5
illustrates, the signals then must pass
through the R1C1 time-constant. This
will reduce the amplitude of low frequency signals, distort square waves
and pulses and can play merry hell
with pulse width modulation (PWM)
signals.
To see why, we need to critically
look at just what it means to feed a
signal through a coupling capacitor.
In Fig.6 we have sketched a PWM
signal which is applied to the left
side of capacitor C1. Below that is the
waveform which appears on the right
hand side of C1 and is displayed on
the oscilloscope screen.
At time t7, the input signal lifts
the left side of C1 from zero to +10V,
charging the capacitor. So the right
side also rises to +10V. Between times
t7 and t8, the input voltage remains
steady. But the charge on C1 leaks
away through the resistor R1, lowering
the voltage on the right hand side of
the capacitor from +10V to +8V. Then
at time t8, the input voltage drops from
+10V to zero.
Because this fall is abrupt, the potential on the right side of the capacitor
must also fall by 10V; ie, from +8V to
Fig.7: one possible circuit
for the Chop/Alternate
section of an analog
scope. CMOS analog
switches alternately
switch the signals from
channels 1 and 2 through
to the vertical deflection
amplifier.
August 1996 69
Fig.8: this series of waveforms illustrates how the Chop mode in an oscilloscope
rapidly chops between the input channels to produce two waveforms on the
screen. Waveforms (c), (d) and (e) show an expansion of the 1ms period in
waveforms (a) and (b).
-2V, taking the displayed trace into
negative regions. This may leak away
to about -1.7V by time t9, when the
input rises again. This time the +10V
change in the input signal lifts the
display up to +8.3V. During the long
constant input between t9 and t10, the
display again leaks down to +6.6V. You
can see that the displayed waveform
is far from the truth.
When the positive input pulses
are long, with a duty cycle greater
than 50%, the display progressively
migrates downwards (duty cycle is
the ratio of the pulse on-time to the
pulse off-time). If the duty cycle remained constant, after many cycles the
displayed signal would be displaced
until the area enclosed between the
positive regions of the waveform and
the zero line is equal to the area enclosed between the negative regions
and the zero line.
70 Silicon Chip
By this rule, the long duty cycle
between times t7 and t12 will push the
waveform downwards. But the same
rule means that between times t13 and
t16, when the duty cycle is short, the
waveform display must rise above the
zero line, in order to equalise positive
and negative areas.
So the complex PWM waveform of
Fig.6 will rise and fall as the duty cycle
changes. The only cure is to monitor
the waveform with DC coupling. AC
coupling is a trap for young players –
use it only when you must block high
DC voltages.
Dual-trace operation
One of the really powerful benefits
of a scope is the ability to monitor two
signals at once but here again there
are traps. If you want to measure the
timing or phase differences between
two signals you need to know just
how your scope displays two different
inputs on the screen simultaneously. What we are talking about is the
choice between Alternate and Chop
modes.
Fig.7(a) illustrates one possible
circuit for the Chop/Alternate section
of an analog scope. Two different signals on channels 1 and 2 firstly pass
through their individual attenuators
and preamplifier stages A1 and A2,
then to the Chop/Alter
nate section
which includes IC1, IC2 and IC3. You
will easily follow its operation as we
view it a bit at a time.
IC1a, b and d are CMOS analog
switches and each turns on only when
a logic high signal is applied to its
control terminal. For example, IC1a
conducts between pins 4 and 3 only
when a logic high is applied to pin 5.
The timebase section of the oscilloscope, as well as providing the
horizontal sweep, also feeds a control
signal in at point T. This controls all
four CMOS switches via inverters
IC2a, IC2b and IC2c. IC3 is a summing
operational amplifier, while Ri1 and
Ri2 are its two input resistors and Rf
is the feedback resistor. Point X is the
summing junction.
The gain from either channel 1
or channel 2 inputs to the output at
point N is -(Rf/Ri) = -(10kΩ/10kΩ) =
-1. Signals from point N feed to the
vertical deflection amplifier for display on screen.
Now what happens when we select
the Alternate display mode? Say we
apply a signal to channel 1 input and
a square wave to channel 2. If the timebase section feeds a low control signal
to the point T, this will be inverted in
IC2b and will present a logic high to
pin 5 of gate IC1a, turning it on.
So the sinewave signal on channel
1 will feed through A1, through Ri1,
IC1a and IC3, and will pass on to the
vertical deflection amplifier, to be displayed on the screen. At the same time,
gate IC1b is off, so channel 2 signals
cannot pass to the vertical deflection
amplifier.
But when a logic high signal is fed to
point T, the conditions reverse. Analog
switches IC1b and IC1c will conduct
and IC1a turn off, allowing the channel
2 signal to be displayed on the screen.
The control signal at point T is high on
the 1st, 3rd, 5th, 7th, etc sweeps and
low on the 2nd, 4th, 6th, 8th, etc. Thus,
all odd sweeps display the sinewave
on channel 1 and all even sweeps show
the square wave on channel 2.
You can use the individual vertical
position (shift) controls to move the
two displays apart. At slow sweep
speeds, the display alternates between
signal 1 in the upper half of the screen
and signal 2 in the lower screen.
At fast sweep speeds, the persistence of the screen phosphor enables
you to see both signals continually on
the screen. Hence, Alternate mode is
successful with fast sweep speeds but
unsuitable at slow sweeps.
Chop mode
Now what happens if you change to
“Chop” mode. This causes a separate
high frequency oscillator within the
timebase unit to toggle the control
signal fed to point T (toggle means to
switch continually between logic high
and low). This is done at a fixed fast
rate, perhaps 10kHz, as illustrated in
Fig.8 but in some high frequency osc
illoscopes the toggle rate may be as
high as 1MHz.
In the example shown in Fig.8, the
main sweep is switched to 100 mil-
liseconds per division, which takes
one second for each full sweep. The
sinewave on channel 1 has a frequency
of about 3.5Hz and the square wave on
channel 2 is about 6Hz. The control
signal at point T has period equal to
1/10kHz = 100µs as Fig.8 shows.
This makes channel 1 conduct
through IC1a for 50µs, channel 2 conducts through IC1b for the next 50µs
and so on. Both input signals are thus
chopped up into thousands of little
time segments 50µs long, like two lines
of ants crawling across a page.
On the screen are displayed these
20,000 discontinuous segments of
the input signals, as IC1a and IC1b
conduct in turn. A small sector of both
traces is shown in Fig.8(d) & (e) drawn
one thousand times time-expanded.
While T is at logic high, a small segment of the sinewave (a) is displayed
in the upper half of the screen. But
when T is at logic low, a short piece
of the square wave (b) appears on the
lower half at (e). While one signal is
displayed, the other is blanked off.
This process continues repeatedly,
right across the screen. The slight
blur
ring due to the width of the
light spot makes each trace appear
continuous.
If we raised the sweep speed sufficiently we would see the discontinuous nature of the display. So chop
mode is unsuitable for very fast sweep
speeds. In some scopes Chop mode is
automatically selected at slow time
base speeds and Alternate is selected
at high sweep speeds.
Now we can see why Chop and
Alternate modes can affect timing
and phase comparisons between two
different signals. Alternate mode leads
to impossibly wrong results, because
it allows the oscilloscope to trigger
independently on each channel; time
correlation is completely lost. Therefore, Chop mode must be used when
comparing the timing of different
signals.
Phase shift
A final vital point to note here is
the phase shift which AC coupling
produces, as noted above. Therefore,
when comparing phases and timing
of different signals, switch both channels to DC coupling or switch both
channels to AC coupling. Don’t have
channel 1 AC-coupled and channel 2
DC-coupled; that will lead to serious
SC
errors.
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regular customer newsletters
BEWARE OF IMITATORS
Direct Importer: AV-COMM PTY. LTD.
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August 1996 71
RADIO CONTROL
BY BOB YOUNG
Multi-channel radio control
transmitter; Pt.7
This month, we deal with the final system
alignment and the programming instructions for
the Mk.22 transmitter. This is mainly a matter of
deciding the features you want and connecting
the various wander leads.
To begin it will be necessary to
re-read the May, June and July 1996
issues of SILICON CHIP in which the
basic instructions for the alignment
of the RF module and the encoder
are discussed. As we left the project
in the July issue, the transmitter was
completely assembled and working up
to the point of radiating a modulated
signal albeit not correctly tuned.
Before we go any further, make sure
your batteries are fully charged before
you start. Charging is accomplished
using a power supply set at 60mA or
a dedicated plugpack charger. These
are available at any good model shop
however you will need to change the
connectors.
The charge plug must be a non-shorting 2.5mm jack type and is inserted
into the socket located on the lower
front right of the Tx. The tip of the
charge plug is wired positive. Remove
all micro-shunts and leads from the
encoder and RF modules.
It is probably best to begin proceedings with the adjustment of the
expanded scale voltmeter circuitry
as this will give a good indication
of the state of your batteries during
the alignment process. Trimpots
VR16 and VR17 on the encoder
PC board control the set points
and range of the meter. VR17 sets
the low point (+8.8V) and VR16
controls the range (sensitivity).
To adjust the meter, hook up
a variable voltage source to the
encoder GND and +9.8V pins on
TB7. Set both trimpots to their
midpoints, set the power supply
to +8.8V and switch on power.
Set the meter pointer to “0” using
VR17. Now increase the voltage
to +10.8V and set the pointer to
“10” using VR16. Drop the volts
back to +8.8V and reset VR17.
Continue this cycle until the
meter reads “10” at +10.8V and
Fig.1: the ideal modulated waveform and
“0” at +8.8V.
recommended rise times.
72 Silicon Chip
With this setting, the meter will peg
immediately after charging and drop
back very quickly to less than “10”
as the surface charge is dissipated.
As nicads are considered exhausted
at 1.1V per cell, the meter will give
an excellent indication of the state of
charge of your batteries. Stop flying at
“0” as you will have only about 10-15
minutes of safe flying after this.
Slip the Tx power input socket
back onto TB7 (PWR) and fire up your
spectrum analyser (yes, as I have stated
before, you will need an analyser) and
plug in the power connector to the RF
module.
Open the May 1996 issue of SILICON
CHIP and work your way through the
tuning sequence presented in that
issue. The production antenna ended
up at 1.3 metres instead of 1.5 metres
long but the tuning range will accommodate that change.
The only area needing special
attention is the final shape of the
modulated waveform. Fig.1 shows the
ideal waveform and recommended rise
Fig.2: an overview of the encoder layout showing the major programming
controls and plug groups. Notice that all eight input configurations are identical.
times. It may be necessary to play with
the value of R7 on the RF module, as
mentioned previously, to adjust for the
spread in the FETs.
Once the RF module is properly
tuned, seal the ferrite slugs with wax
to prevent them moving. It is now
possible to drive a receiver from the
transmitter. As all input stages have
been disabled, only the default waveform will be transmitted (all 1.5ms
pulses). Switch on the receiver and
plug a servo set to 1.5ms neutral (most
modern servos) into channel 1.
Better still, plug in a pulse width
meter. Remove all leads and micro-shunts from the encoder PC board,
switch on the transmitter and the pulse
width meter should read 1.5ms or the
servo move to neutral, if you have
followed the instructions in the June
1996 issue.
The 10kΩ 10-turn trimpot VR2
(NEUT) is there to provide neutral adjustment. Clockwise rotation increases
the pulse width. Use this to set the
neutral if it is not already correct and
switch the Tx OFF. Try to get into the
habit of changing the plugs and sockets
with the Tx OFF. There are only one
or two plugs that may cause problems
and these are not usually moved once
in place (power sockets). The Tx must
be switched OFF when changing the
RF module.
The Mk.22 transmitter is now ready
for business, so let us move on to the
real work.
Fig.3: each channel input has
three main components: the
VARY-NORMAL 3-pin header
set, the CHANNEL INPUT 3-pin
header set and an Adjustable
Servo Travel Volume (ATV)
potentiometer.
(see Fig.2, page 80, July 1996 issue)
which are free to wander anywhere
on the PC board. All control elements
are wired in an identical fashion with
the centre lead carrying the signal and
the two outside leads for positive and
negative. All control inputs on the encoder are fitted with identical, mating
3-pin headers (plugs). Any control
may be connected to any channel in
any order.
This arrangement results in a transmitter of the utmost flexibility. Even
the front panel controls can be programmed in the most suitable manner
for the task at hand. Toggle switches
can become retract switches, dual rate
switches or mix IN-OUT switches.
Sense of operation may be reversed,
channel allocation changed or direction of the servo travel reversed very
quickly and without complex menu
stepping.
Fig.2 gives an overview of the
encoder layout showing the major
programming controls and plug
groups. Notice that all eight input
configurations are identical so that
it is only necessary to master one to
have complete mastery over all eight
or indeed 24 channels.
Each channel input has three main
components: the VARY-NORMAL
3-pin header set, the CHANNEL INPUT
3-pin header set and an Adjustable
Servo Travel Volume (ATV) potentiometer – see Fig.3. Each of the channel input sets are numbered on the PC
board and run from left to right. These
three items give rise to an almost limitless variety of programming options.
We will work through some of these
options, paying particular attention to
the basic principles involved, in order
to build a good knowledge of how the
system works. This will make the more
complex programming tasks (such as
CROW) much easier to understand
when we describe them in coming
issues.
Programming the encoder
The Mk.22 encoder utilises what
is perhaps best defined as “Wander
Lead” programming. All controls are
wired with identical 3-pin sockets
Fig.4: this diagram shows how the various micro-shunts (shorting
links) must be placed across TB10, if the configuration module is
not used.
August 1996 73
We will begin with the simplest and
most fundamental tasks and work forward from there. Load the appropriate
micro-shunts and sockets as we go
through the programming sequence.
Configuration module
The configuration module is not
used in the basic Mk.22 transmitter.
This was briefly mentioned and pictured in the June 1996 issue. If the
module is not plugged into the configuration port TB10, then micro-shunts
(shorting links) must be placed across
TB10 as shown in Fig.4 to complete
these open circuit input leads (see
circuit in the March 1996 issue). These
micro-shunts also play an important
role in the mixing programming and
we will deal with that later.
Channel allocation
As a result of the wander lead concept, channel allocation is a matter of
deciding which controls should utilise
which channel and plugging the 3-pin
sockets onto the appropriate CHANNEL INPUT header pins. Channel
allocation is a most important function
when we come to such complex programming options as CROW, changing
stick modes or matching a Mk.22 Tx
to another brand of radio. As glitches
tend to affect channel 1 more than any
other channel, it is best to keep the
flying controls away from channel 1.
The standard Silvertone channel
allocation is as follows:
Channel
1
Allocation
motor
2
aileron
3
elevator
4
rudder
5
gear
6
flaps
7
aux 1
8
aux 2
Other brands of R/C equipment
use different channel allocations. To
match a Mk.22 Tx to a model already
fitted with another brand of receiver,
it is a simple matter to duplicate the
channel allocation by rearranging the
order of the sockets.
Servo reversing
Servo reversing is simply a matter
of rotating the 3-pin socket on any
74 Silicon Chip
and we will now move on to some of
the more advanced features.
End point adjustment
Fig.5: the sense of operation on the
toggle switch for dual rate operation
can be reversed by reversing the 3-pin
socket on the VARY-NORMAL header.
Adjusting throttle linkages can be
a tricky business as it is often almost
impossible to get both ends exactly
right. Using the ATV in conjunction
with adjustable linkages overcomes
this problem. Whilst not a true end
point adjustment it certainly will adjust both end points simultaneously
and will set the exact amount of servo
travel needed to match the carburettor
arm travel.
Fig.6: use this diagram when pro
gramming toggle switch operation.
Dual rate programming
of the CONTROL INPUT headers by
180°. Keep in mind that any error
from absolute neutral will be doubled.
For example, if the throttle servo is at
one end before reversing, then it will
immediately fly to the other end when
reversed. If the trim is at absolute neutral when the socket is reversed, then
no servo movement will be apparent.
Programming servo travel
In order to simplify the programming, certain configurations call for
the ATV (adjustable servo travel volume) potentiometer to be connected to
become a completely different type of
volume control. The VARY-NORMAL
header pin sets provide this function.
When a micro-shunt is placed on the
centre/left pair of pins as in Fig.5(a),
the CHANNEL GAIN potentiometer is
programmed as the ATV potentiometer. In this mode, servo travel may be
adjusted from 20-120% (0.9 - 2.1ms)
of the normal servo travel using the
channel gain (ATV) potentiometer.
Clockwise rotation increases the
amount of servo travel.
If the micro-shunt is placed on the
centre/right pair of header pins, as in
Fig.5(b), then the ATV pot is allocated
to other functions and the servo travel
reverts to the NORMAL non-adjustable
100% level (1 - 2ms) and the ATV potentiometer is no longer available for
servo travel adjustment.
At this point the transmitter should
have all of the micro-shunts loaded
on TB10, the main controls hooked
to the CHANNEL INPUT headers
and the micro-shunts loaded on the
VARY-NORMAL headers. You can now
move multiple servos simultaneously.
This completes the basic programming
All 24 channels may be programmed
for DUAL RATE operation. Simply
remove the micro-shunt from the
VARY-NORMAL headers on the channels intended for DUAL RATE operation and connect the desired toggle
switches to the appropriate headers.
Any of the front panel toggles can
be used on any channel. The choice
should be based on convenience of
operation.
Sense of operation of the toggle
switch can again be reversed by simply reversing the 3-pin socket on the
VARY-NORMAL header – see Fig.5.
When the toggle is in the VARY position the ATV potentiometer becomes
the DUAL RATE set pot. Thus with the
switch in the NORMAL location, the
ATV pot is disabled and a non-adjustable 100% servo travel is available.
With the switch in the VARY position,
the ATV pot is used to set the amount
of DUAL RATE variation. It is usual
to select the VARY position with the
toggle DOWN.
The amount of DUAL RATE adjustment ranges from 20% to 120%.
Anticlockwise rotation decreases the
amount of servo travel. Note that it is
possible to program the Mk.22 Tx for
increased throw in the DUAL RATE
setting.
Toggle switch programming
There are two identical toggle switch
modules built into the main encoder
PC board module. These are located
on the righthand side of the PC board
just above the righthand input groups
(see Fig.2) These modules consist of
two 3-pin headers and a potentio
meter.
Fig.6 shows these in detail. Note
that the lefthand 3-pin header of each
toggle module is labelled SW and this
Kit Availability
Kits for the Mk.22 transmitter are available in several different forms, as follows:
Fully assembled transmitter module......................................................$125.00
Basic transmitter kit (less crystal)............................................................$89.00
Transmitter PC board...............................................................................$29.50
Crystal (29MHz).........................................................................................$8.50
Fully assembled encoder module..........................................................$159.00
Encoder kit.............................................................................................$110.00
Encoder PC board...................................................................................$29.50
Transmitter case kit................................................................................$395.00
Full transmitter kit (includes all the above).............................................$594.00
Post and packing of the above kits is $3.00. Payment may be made by Bankcard,
cheque or money order to Silvertone Electronics, PO Box 580, Riverwood, NSW
2210. Phone (02) 533 3517.
will receive the 3-pin socket from the
toggle switch (see Fig.2c, July 1996
issue). Thus, any toggle switch on
the Tx front panel may be used as the
actuator for the toggle channels.
The righthand 3-pin header labelled
CH is the output connection. The short
jumper cable with a 3-pin socket at
each end (see Fig.2e, July 1996 issue)
is connected to this header. The other
end of this patch cord can go to any
CHANNEL INPUT header in your
channel allocation plan. Thus any two
channels may be allocated to toggle
switch actuation.
Servo reversing is available simply
by reversing the CHANNEL INPUT
socket as normal. A novel feature is
the ability to very quickly reverse the
sense of operation of the toggle switch
by simply reversing the socket on the
SW header. Thus UP-ON becomes
DOWN-ON.
With the channel input programmed
for NORMAL mode, adjust
ing the
toggle module potentiometer will
provide from almost zero to 100%
travel volume. Clockwise increases
the servo travel.
Another novel feature of this arrangement is that 180° of servo travel
is easily obtained by using the toggle
module potentiometer in conjunction
with the VARY mode ATV pot. Some
care is needed here in case the brand
of servo you are using has its rotation
angle limited by internal stops to
less than 180°. Check to ensure that
the servo is not straining against the
internal end stops.
There is provision for two toggle
modules on the standard encoder PC
board.
Programming knob control
The standard Mk.22 case is punched
for four toggle switches and two knob
controls. The knob control consists of
a panel-mount potentiometer upon
which are mounted limiting resistors
and a cable fitted with a 3-pin socket
(see Fig.2d, July 1996). The resistors
allow the full 270° of rotation to be
used without driving the channel beyond the electronic limits allowable.
Thus, the knob control is a completely self-contained proportional
control element which may be treated
as one axis of a 2-axis stick assembly.
It may be allocated and reversed in the
normal manner.
There are two knob controls in the
standard Mk.22 transmitter, however
all eight channels could easily be knob
controls in a suitable case.
Programming slide control
A slide control unit is available as
an option and again may be considered
a single axis proportional control ele
ment. It may be allocated and reversed
as normal. However, this would require a slot to be cut in the case by
hand. A slide control suitable for flaps
is available as an optional extra.
That is all that space allows for this
month. Next month we will discuss
mixing, dual control and frequency
SC
interlock.
Scan Audio Pty Ltd
August 1996 75
An introduction to
IGBTs
When it comes to high power switching applications
circuit designers generally choose between bipolar
transistors or Mosfets. But there is an alternative
which combines the best of both devices – the insulated
gate bipolar transistor or IGBT. It can be thought as a
bipolar transistor with a high impedance gate instead
of a low impedance base.
More and more we are seeing heavy duty switchmode power circuits – inverters, power supplies, induction motor control and so on. As the applications
continue to become more stringent, semiconductor
manufacturers need to create products that approach
the ideal switch. The ideal switch would have: (1)
zero resistance or forward voltage drop in the on-state;
(2) infinite resistance in the off-state; (3) switch on
and off with infinite speed; and (4) would not require
any input power to make it switch.
Fig.1: reduced forward voltage drop of an IGBT compared
to a Mosfet with similar ratings.
76 Silicon Chip
Since we don’t yet have the ideal switch, designers
must choose a device that best suits the application.
The choice involves considerations such as voltage,
current, switching speed, drive circuitry, load and temperature effects. There are a variety of solid state switch
types available and they all have their strong and weak
points.
High voltage power Mosfets
The characteristics that are most desirable in a solid-state switch are fast switching speed, simple drive
requirements and low conduction loss.
For low voltage applications, power Mosfets offer
very low on-resistance [RDS(on)] and approach the
desired ideal switch. But in high voltage applications,
Mosfets exhibit increased RDS(on) which results in
increased conduction losses. In a power Mosfet, the
on-resistance is proportional to the breakdown voltage
raised to approximately 2.7:
RDS(on) = (VDS)2.7
Mosfet technology has now advanced to a point
where RDS(on) is near the theoretical limit. A new
approach is needed to obtain very low on-resistance
without sacrificing switching speed. This is where the
IGBT comes in.
By combining the low conduction loss of a BJT
(bipolar junction transistor) with the switching speed
of a power Mosfet an optimal solid state switch would
be obtained. In fact, the IGBT is a spin-off from power
Mosfet technology and its structure closely resembles
Fig.2: reduced die size of an IGBT compared to a Mosfet
with similar ratings.
Fig.3: reduced package size of an IGBT compared to a
Mosfet with similar ratings.
that of a power Mosfet. The IGBT has a high input
impedance and fast turn-on like a Mosfet. And they
have an on-voltage and current density comparable to a
bipolar transistor.
Compared to SCRs, the IGBT is faster, has better dv/dt
immunity and above all, has better gate turn-off capability. While GTOs (gate turn-off SCRs) are capable of
being turned off at the gate, substantial reverse gate
current is required, whereas turning off an IGBT only
requires the gate capacitance to be discharged. Against
that, SCRs have a slightly lower forward voltage and a
higher surge current capability than IGBTs.
Many of today’s switching circuits use Mosfets
because of their simple gate drive. Since the structure
of both devices is similar, the change to IGBTs can be
made without having to redesign the gate drive circuit.
Like Mosfets, IGBTs are transconductance devices and
can remain fully on if the gate voltage is held above a
certain threshold.
As shown in Fig.1, using an IGBT in place of a
power Mosfet dramatically reduces the forward voltage drop at currents above 12 amps. By reducing the
forward drop, the conduction loss is decreased. The
gradual rising slope of the Mosfet in Fig.1 can be
attributed to the relationship of VDS to RDS(on). The
IGBT curve has an offset due to an internal forward
biased p-n junction and a fast rising slope typical of a
minority carrier device.
Replacing a Mosfet with an IGBT can improve the
efficiency and/or reduce the cost. As shown in Fig.2, an
IGBT has considerably less silicon area than a similarly
rated Mosfet. The reduced silicon area makes the IGBT
the lower cost solution. Fig.3 shows the package area
reduction by using an IGBT. This suits it for designs
where space is restricted.
Speaking IGBT
Before we go any further, perhaps we should tell
you how to say IGBT. Instead of referring to them as
“Iggbets” most designers call them by the initials,
“eye gee bee tees” – more of a mouthful perhaps but
that’s the way it is.
IGBTs are replacing Mosfets in high voltage applications where conduction losses must be kept low. In
fact, SILICON CHIP featured a 2kW sinewave inverter
with IGBTs in the October 1992 to February 1993
issues. Four 1kV IGBTs were used in the high voltage
H-pack section where 365V DC is converted to a 50Hz
sinewave using pulse width modulation at around
4kHz. In this instance, we were forced to use IGBTs
because no combination of currently available power
Mosfets was sufficiently rugged for the job.
With zero current switching or resonant switching
techniques, IGBTs can be operated in the hundreds of
kilohertz range. Typically though, although turn-on
speeds are very fast, turn-off of the IGBT is slower than
a Mosfet. It exhibits a significant current fall time or
“tailing”. This tailing restricts IGBTs to operating at less
than 50kHz in traditional “square wave” PWM switching applications.
Up to 50kHz then, IGBTs are often a better solution
than bipolar transistors, Mosfets or thyristors (SCRs).
Fig.4: forward voltage
drop (VCE(sat)) and fall
time (tf) has improved
since IGBTs were
introduced.
August 1996 77
Introduction to IGBTs – continued
Fig.5: cross-section and equivalent schematic of an
insulated gate bipolar transistor (IGBT) cell.
When compared to BJTs, IGBTs have similar ratings
in terms of voltage and current but the isolated gate in
an IGBT makes it simpler to drive. BJTs used as switches require sufficient base current to maintain saturation.
Typically, the base current needs to be at least 1/10th of
the collector current. BJT drive circuits must therefore
be sensitive to variable load conditions.
In other words, base current for a BJT must be kept
proportional to the collector current; otherwise the device will come out of saturation with high-current loads
and will have excessive base drive under low-load
conditions. Either way, it can lead to increased power
dissipation.
BJTs are minority carrier devices and charge storage
Fig.6: cross-section and equivalent schematic of a metaloxide-semiconductor field-effect transistor (Mosfet) cell.
78 Silicon Chip
effects including recombination slow the performance
when compared to majority carrier devices such as
Mosfets. IGBTs also experience recombination that
accounts for the current “tailing”, yet IGBTs have been
observed to switch faster than BJTs.
Since the introduction of IGBTs in the early 1980s,
semiconductor manufacturers have learned how to
make the devices faster. As illustrated in Fig.4, some
trade-offs in conduction loss versus switching speed
exist. Lower frequency applications can tolerate slower
switching devices. Because the loss period is a small
percentage of the total on-time, slower switching is
traded for lower conduction loss. In a higher frequency
application, just the opposite would be true and the
device would be made faster and have greater conduction losses.
Notice that the curves in Fig.4 show reductions in
both the forward drop VCE(sat) and the fall time tf of
newer generation devices. These capabilities suit the
IGBT for applications such as motor control, power
supplies and inverters which require devices rated at
600-1200V.
IGBT structure
The structure of an IGBT is similar to that of a double diffused (DMOS) power Mosfet. One difference
between a Mosfet and an IGBT is
the substrate of the starting material.
By varying the starting material and
altering certain process steps, an
IGBT may be produced from a power
Mosfet mask; however, at Motorola,
mask sets are designed specifically
for IGBTs. In a Mosfet the substance
is P+ as shown in Fig.5.
The n- epi resistivity determines
the breakdown voltage of a Mosfet as
mentioned earlier using the relationship: RDS(on) = (VDS)2.7
To increase the breakdown voltFig.7: the
age of the Mosfet, the n- epi region
symbols for
thickness (vertical direction in the
IGBTs (a) and
diagram) is increased. Reducing
Mosfets (b).
the RDS(on) of a high voltage device
requires a greater silicon area to make up for the increased n- epi region.
The effects of the high resistive n-epi region were
overcome by conductivity modulation. The n-epi was
placed on the P+ substrate, forming a pn junction
where conductivity modulation takes place. Because of
conductivity modulation, the IGBT has a much greater
current density than a power Mosfet and the forward
voltage drop is reduced. Now the P+ substrate, n-epi
layer and P+ “emitter” form a BJT transistor and the
n-epi acts as a wide base region.
Current tailing has been mentioned above. The device structure shown in Fig.5 provides an insight into
tailing. Minority carriers build up to form the basis for
conductivity modulation. When the IGBT turns off,
these carriers do not have a current path to exit the
device. Recombination is the only way to eliminate
the stored charge resulting from the build-up of excess
carriers. Additional recombination centres are formed
Fig.8: IGBT current turn-off waveform.
by placing an N+ buffer layer between the n-epi and P+
substrate.
While the N+ buffer layer may speed up recombination, it also increases the forward voltage drop. Hence
the tradeoff between switching speed and conduction
loss becomes a factor in optimising performance. The
N+ buffer layer also prevents thermal runaway and
punch-through of the depletion region. This allows a
thinner n- epi to be used which somewhat decreases
forward voltage drop.
Four layers
The IGBT has a four layer (PNPN) structure, resembling that of an SCR. But unlike the SCR where the
device latches on and gate control is lost, an IGBT is
designed so that it does not latch on. Full gate control is
available at all times.
Because the IGBT is a four-layer structure, it does not
have the inverse parallel diode inherent in power Mosfets. This is a disadvantage to motor control designers
who use the anti-parallel diode to recover energy from
the motor.
Like a Mosfet, the gate of an IGBT is electrically isolated from the rest of the chip by a thin layer of silicon
dioxide, SiO2. This gives it a high input impedance and
excellent drive efficiency.
a voltage across the base-emitter junction of the NPN.
If the base-emitter voltage is above a certain threshold
level, the NPN will begin to conduct causing the NPN
and PNP to enhance each other’s current flow and both
devices can become saturated. This results in the device
latching on in a fashion similar to an SCR. Device pro
cessing directs currents within the device and keeps the
voltage across Rshorting low to avoid latching.
The IGBT can be gated off, unlike the SCR which has
to wait for the current to cease, allowing recombination
to take place in order to turn off. IGBTs offer an advantage over the SCR by controlling the current with the
device, not the device with the current. The internal
Mosfet of the IGBT when gated off will stop current
flow and at that point, the stored charges can only be
dissipated through recombination.
The IGBT’s on-voltage is represented by the sum of
the offset voltage of the collector base junction of the
PNP transistor, the voltage drop across the modulated resistance Rmod and the channel resistance of the
internal Mosfet. Unlike the Mosfet where increased
temperature results in increased RDS(on) and increased
forward voltage drop, the forward drop of an IGBT stays
relatively unchanged at increased temperatures.
Switching speed
Until recently, slow turn-off speed limited IGBTs
from serving a wide variety of applications. While
turn-on is fairly rapid, initial IGBTs had current fall
times of around three microseconds. The turn-off time
of an IGBT is slow because many minority carriers are
stored in the n- epi region. When the gate is initially
brought below the threshold voltage, the n- epi contains a very large concentration of electrons and there
will be significant injection into the P+ substrate and
a corresponding hole injection into the n- epi. As the
electron concentration in the n-region decreases, electron injection decreases, leaving the rest of the electrons
to recombine.
Therefore, the turn-off of an IGBT has two phases: an
injection phase where the collector current falls very
Equivalent circuit
IGBT operation is best understood by again referring
to the cross section of the device and its equivalent
circuit shown in Fig.5. Current flowing from collector
to emitter must pass through a pn junction formed by
the P+ substrate and n- epi layer. This drop is similar
to that seen in a forward biased pn junction diode and
results in an offset voltage in the output characteristic.
Current flow contributions are shown in Fig.5 using
varying line thickness, with the thicker lines indicating
a high current path. For a fast device, the N+ buffer
layer is highly doped for recombination and speedy
turn off. The additional doping keeps the gain of the
PNP low and allows two-thirds of the current to flow
through the base of the PNP (electron current) while
one-third passes through the collector (hole current).
Rshorting is the parasitic resistance of the P+ emitter
region. Current flowing through Rshorting can result in
Fig.9: cross-section and equivalent schematic of a short
circuit rated IGBT cell.
August 1996 79
Introduction to IGBTs – continued
quickly and a recombination phase in which the collector current decreases more slowly.
Fig.8 shows the switching waveform and the contributing factors to tail time of a “fast” IGBT designed for
PWM motor control. In power Mosfets, the switching
speed can be greatly affected by the impedance in the
gate drive circuit and the same rules apply to IGBTs.
Comparing IGBTs, BJTs & Mosfets
The conduction loss of BJTs and IGBTs is related to
the forward voltage drop of the device while a Mosfet’s
conduction loss is based on RDS(on). Table 1 gives a comparison of turn-off and conduction losses at 10 amps for
a power Mosfet, an IGBT and a BJT at junction temperatures of 25°C and 150°C.
Note that while the devices in Table 1 have approximately the same ratings, their chip sizes vary significantly. The bipolar transistor requires 1.2 times more
silicon area than the IGBT while the Mosfet requires 2.2
times the area of the IGBT. This difference in die area
has a direct effect on the cost of the devices.
At higher currents and high temperatures, the IGBT
offers low forward drop and a switching time similar to
the BJT without the drive difficulties. The lower conduction losses of the IGBT reduce power dissipation
and heatsink size.
Thermal resistance
An IGBT and power Mosfet produced from the same
size die have similar junction-to-case thermal resistance
Fig.11: IGBTs offer performance advantages in PWM
variable-speed induction motor drives. They can directly
control 3-phase motors from a rectified mains supply.
80 Silicon Chip
Fig.10: the waveforms associated with anti-parallel diode
turn-off.
because of their similar structures.
Short circuit rated devices
Using IGBTs in motor control circuit requires them to
withstand short circuit current for a given period. Although this varies with the application, a typical value
of ten microseconds is used for designing these specialised IGBTs. Notice that this is only a typical value given
on the data sheet. IGBTs can be made to withstand short
circuit conditions by altering the device structure to include an additional resistance (Re, in Fig.9) in the main
current path.
The benefits associated with the additional series
resistance are twofold.
First, the voltage created across
Characteristic
Re, by the large current passing
through Re, increases the percentCurrent Rating
age of the gate voltage across Re, by
Voltage Rating
the classic voltage divider equation.
R(on) <at> TJ = 25°C
Assuming the drive voltage applied
to the gate-to-emitter remains the
R(on) <at> TJ = 150°C
same, the voltage actually applied
Fall Time (typical)
across the gate-to-source portion of
* Indicates VCEO rating
the device is now lower. This causes the device to operate in an area
of the transconductance curve that
reduces the gain and it will pass less current.
Second, the voltage developed across Re results in a
similar division of voltage across Rshorting and VBE of
the NPN transistor. The NPN will be less likely to attain
a VBE high enough to turn the device on and cause a
latch-up situation.
These two situations work together to protect the device from catastrophic failure. The protection period is
specified in the ratings, giving the circuit time to detect
a fault and shut off the device.
The introduction of the series resistance Re also
results in additional power loss by slightly elevating the
forward drop of the device. However, the magnitude of
short circuit current is large enough to require a very
low Re value. The additional conduction loss of the
device due to the presence of Re is not excessive when
comparing a short-circuit rated IGBT to a non-short
circuit rated device.
Anti-parallel diode
When using IGBTs for motor control, designers have
to place a diode in anti-parallel across each device in
order to handle the regenerative or inductive currents
of the motor. The optimal setup is to have the diode
co-packaged with the device. A specific line of IGBTs
has been created by Motorola to address this issue.
These devices work very well in applications where
energy is recovered to the source and are favoured by
Table 1: Device Characteristics
TMOS
IGBT
Bipolar
20A
20A
20A
500V
600V
500V*
0.2 ohms
0.24 ohms
0.18 ohms
0.6 ohms
0.23 ohms
0.24 ohms**
40ns
200ns
200ns
** BJT TJ = 100°C
motor control designers.
Like the switching device itself, the anti-parallel
diode should exhibit low leakage current, low forward
voltage drop and fast switching speed. As shown in
Fig.10, the diode forward drop multiplied by the average current it passes is the total conduction loss produced. In addition, large reverse recovery currents can
escalate switching losses.
A secondary effect caused by large reverse recovery
currents is EMI at the switching frequency and the
frequency of the resulting ringing waveform. This EMI
requires additional filtering in the circuit. By co-packaging the IGBT with its anti-parallel diode, the parasitic inductances that contribute to ringing are greatly
reduced.
Induction motor drive
Mains operated, PWM variable speed motor drives
are an application well suited for IGBTs. As shown
in Fig.11, IGBTs may be used to directly control the
voltage supplied to a 3-phase motor to control its
speed. Depending on the application, the IGBT may be
required to operate from the full-wave rectified mains
supply.
Acknowledgement
This article reproduced by arrangement from Motorola Semiconductor Application Note AN1541.
SC
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Model KSN 1141
The new Powerline series of Motorola’s
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This results in a product that is practically
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Frequency Response: 1.8kHz - 30kHz
Av. Sens: 92dB <at> 1m/2.83v (1 watt <at> 8Ω)
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Max. Temperature: 80°C
Typ. Imp: appears as a 0.3µF capacitor
Typical Frequency Response
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August 1996 81
Creating shortcuts on the desktop
1: right click the item and drag it onto the desktop.
2: click “Create Shortcut Here” from the menu.
Customising the Win95
Computer Bits
desktop & start menus
The Windows 3.11 Program Manager
is obsolete. In its place, Windows 95
presents a slick new interface that lets
you place icons and folders directly on
the desktop. Here’s how to go about it.
By GREG SWAIN
Unlike its predecessor, the Win95
desktop can play host to virtually anything you care to drag there. While
the Windows 3.11 desktop limits you
to the Program Manager and the icons
of minimised applications, the Win95
desktop is a far friendlier place to be.
Want to place shortcuts to your
drives directly on the desktop? No
problem – just open My Computer,
right click (yes, right click) the relevant drive, drag it onto the desktop
and release the mouse button. Choose
82 Silicon Chip
“Create Shortcut Here” when the popup menu appears and there’s your
shortcut.
If you now double-click on the new
shortcut icon, the Explorer opens to
show the contents of the drive.
You can do exactly the same thing to
folders (the new word for directories),
applications or even individual files.
All you have to do is launch the Explorer, right click on the appropriate
folder, executable (exe) file or program
file, and drag it onto the desktop. When
the job is done,
the application’s
icon appears on the
desktop but with
one minor difference – there’s a little
arrow to indicate
that it is a short
cut to the application (see example
at right).
Don’t clutter your desktop with
shortcuts though. They should be
reserved for your most frequently
used appli
c ations. When you no
longer want a particular shortcut on
the desktop, just drag it to the Recycle Bin. Note that this gets rid of the
shortcut only and not the original file
or hardware item.
Renaming shortcuts
When you create a shortcut, Win95
automatically adds the words “Short
cut to” to the desktop icon; eg, “Short
cut to Explorer”. However, the little
3: that’s it – your shortcut appears on the desktop.
4: right-click the shortcut icon to rename it.
Rearranging the start menus
Problem: the CD Player entry is buried four menus deep.
1: right click the Start button, then left click “Explore”.
2: the Explorer opens at the Start Menu folder.
3: “drill” down to the Multimedia folder.
continued next page
August 1996 83
4: left-click the CD Player shortcut and drag it onto
the Start Menu folder.
arrow that’s added to the icon makes
it obvious that it’s a shortcut so these
words are superfluous. Deleting them
is easy – just right click on the icon
and left click on “Rename” from the
pop-up menu. It’s now simply a matter of typing in the new name for the
shortcut and left clicking off the icon.
By the way, get used to using the
right mouse button when you install
Windows 95. Right click on just about
anything, including the Task Bar, and
a menu pops up that lets you carry out
certain functions. Unlike Windows
3.11, the right mouse button now actually does something useful and it’s
easy to use.
Rearranging the Start menus
Apart from using desktop shortcuts,
applications are usually launched via
5: now when you click the Start button, the CD Player
entry appears in the first menu.
the Start button. When you click the
Start but
ton, you navigate through
a series of menus to the application
you want.
Windows 95 automatically adds its
own applications to the Start menus
during installation. Any applications
that you later install are also automatically added and these can even
include entries for readme files or
on-line registration of the software.
As a result, your Start menus quickly become cluttered with entries that
are seldom (if ever) used. Worse still,
an application that you use frequently
can be buried three or four menus
deep and drilling down to it each
time you want to run it can become
annoying.
Fortunately, it’s easy to rearrange
the Start menus to suit the way you
Moving the status bar
You can move the
Task Bar to the top
of the screen by
left clicking on it
and dragging it to
its new location.
It automatically
snaps into place
and the desktop
icons move to
make room for it.
84 Silicon Chip
want to work. The first thing to realise here is that the entries in the Start
menus mirror the entries in the Start
Menu folder and its sub-folders when
you open the Explorer. The second
thing to realise is that these entries
are shortcuts and not the actual files
themselves, as indicated by the little
arrows attached to their icons.
What’s the easiest way of getting to
the Start Menu folder in the Explorer?
Just right click the Start button and
then choose “Explore” from the popup menu. From there, you can start
explor
ing the contents of the Start
Menu folder and its sub-folders.
To delete an entry (eg, a readme
shortcut), just drag it from the Explorer to the Recycle Bin. To move
an entry, just left click it and drag it
to its new location. Now, when you
want to launch the application via
the Start button, it will appear on the
corresponding menu.
In the example given, the CD Player was buried four menus deep. By
opening the Explorer and dragging
the Shortcut entry directly to the Start
Menu folder, it now appears on the
opening menu. You can do this to any
individual item or to groups of items.
Moving the status bar
Finally, if you don’t like having
the Task Bar along the bottom of the
screen, left click it with the mouse
and move it. It can be “snapped” into
position along one side of the screen
or along the top and your desktop
icons will automatically adjust their
SC
positions to make room for it.
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battery is not supplied: $35, suitable plugpack for
the receiver: $10 ...NEW - LOW COST 2 CHANNEL
UHF REMOTE CONTROL: Two channel encoded UHF
remote control has a small keyring style assembled
transmitter, kit receiver has 5A relay contact output,
can be arranged for toggle or momentary operation:
$35 for one Tx and one Rx, additional Tx’s $12 Ea.
OATLEY ELECTRONICS
PO Box 89
Oatley NSW 2223
Phone (02) 9584 3563
Fax (02) 9584 3561
orders by e-mail:
branko<at>oatleyelectronics.com
major cards with phone and fax orders,
P&P typically $6.
VINTAGE RADIO
By JOHN HILL
A rummage through my junk
Although I have been to many swap meets of
various kinds, the last one I attended was special because it was a vintage radio swap meet.
It was held in Melbourne and was very well
attended. What’s more, I had taken a site at the
meet for the express purpose of selling some of
my junk.
There is one serious problem associated with collecting and that is the
slow but steady accumulation of bits
and pieces over the years. As most
readers would know, some of these bits
and pieces are extremely valuable and
supply restorers with many otherwise
unobtainable spares. But it gets out of
hand after a while, so I decided to be
ruthless and off-load some of my junk
so that I could to take possession of my
shed again. The swap meet seemed like
an appropriate place for its disposal.
Sifting through the rubble was great
fun and all sorts of things were found
that had been completely forgotten.
When sort
ing through these treasures, it was initially a case of “no, I
shouldn’t sell that”, or “I must keep
those”, or “these may come in handy”,
and so on. So, by the end of the day,
hardly a thing had been set aside for
the big sale. As a result, the process
had to be repeated with a little more
resolve.
My scrounging uncovered a few
interesting relics. As some which I
earmarked to sell are fairly rare and
This old magnetic pick-up was made to fit straight onto the sound arm of an
acoustic phonograph.
86 Silicon Chip
likely be of inter
est to readers, it
seemed like a good opportunity to
photograph them and write them up
for Vintage Radio. Even though these
things are quite collectable, I had no
real use for them and the larger items
were only taking up valuable space.
The first of these interesting items
is a magnetic pick-up head for 78rpm
recordings. This particular pick-up
was specially made to fit onto the tone
arm of an acoustic phonograph, thus
allowing records to be played through
a radio receiver.
While playing the family phonograph through a radio was common
practice in the late 1920s and early
1930s, it was usually done using a
complete pick-up with an accompany
ing volume control. A pick-up head
only that fitted on to the phonograph’s
tone arm would have been a less
expensive option. However, its very
long, unshielded lead to the receiver
may have caused some hum problems.
Battery eliminator
The next item is from 1927 and is
a “B” battery eliminator. These units
were usually large and heavy and this
Australian-made Emmco was no exception. It uses a cold cathode rectifier
and supplies a range of “B” voltages
only. Some eliminators incorporated
“C” voltages as well.
While the “B” battery eliminator
solved the expense of frequent “B”
battery purchases, the rechargeable
lead acid “A” battery was another
problem in that it required recharging
at regular intervals, which was fairly
inconvenient.
Shown in one of the accompanying
photographs is a Philips “A” battery
trickle charger. Its job was to slowly
and continu
ously recharge the “A”
battery – hopefully at a rate which
This photo shows an Emmco “B” battery eliminator.
It used a cold cathode rectifier and had three output
voltages, two of which could be varied using the large
knobs on the top of the unit.
approximated the discharge rate/period – and eliminate the irksome task
of carting the battery off to the nearest
garage or battery service centre.
Of course, neither the “B” battery
eliminator nor the “A” battery trickle
charger were of any use unless 240V
AC power was available. Back in the
1920s, only the cities and larger towns
had AC electric power and out in the
country, beyond these supply systems,
receivers still used batteries, just as
they had done since radio first began.
Radio had not been with us long
when someone reckoned that having
one in their car would be a great idea.
The vibrator unit was the big breakthrough in battery powered receivers
because it allowed a radio to operate
on a single battery – usually a 6V or
12V lead acid type. A vibrator, in
conjunction with a special transformer
and a rectifier valve, was the heart of
car radio receivers up until about 1960.
But there were a few car radios
before the vibrator came on the
scene. These receivers still required
a high tension supply and it was obtained from a motor/generator set (a
low-voltage electric motor driving a
high-voltage generator). These devices
produced quite high voltages – up to
180V in the case of the Emerson unit
shown in one of the accompanying
photographs.
No doubt the engineering involved
in manufacturing a motor/generator
was considerable and its cost was
probably equal to that of the receiver
itself.
It is amazing how many ingenious
A Philips “A” battery trickle charger. The rectifier valve
(right) plugs into the large hole at top right, while the two
smaller holes are for the battery leads. Power is applied to
the socket on the left.
and well designed products appeared
in the early days of radio, only to be
rendered totally obsolete in a very
short time. The car radio motor/generator unit would be a classic example of
instant obsolescence once the vibrator
arrived on the scene.
(Editorial comment: although the
motor generator had only a short
life in car radio applications, larger
versions were used extensively by
the armed forces during World War
ll and beyond – until the end of the
valve era, in fact. They were woefully
inefficient devices. One of the top
brands, the “Genemotor”, could boast
an efficiency of only 30% but this was
not regarded as a serious problem for
military applications).
4-gang capacitor
Shown in one of the photographs is
a 4-gang tuning capacitor from an ancient TRF receiver. After the superhet
became established, tuning capacitors
were mainly two and 3-gang types but
some of the old TRF capacitors were
four and 5-gang units.
This old 4-gang capacitor is quite
large, as was the norm back then, and
is made entirely of brass. Finding a
practical use for such a monstrosity is
This elaborate device was used to power early car radios. Made by Emerson, it
consists of a low voltage DC electric motor driving a high voltage DC generator.
It was capable of producing 180V at 80mA. The advent of the vibrator rendered
these monstrous things obsolete for car radios.
August 1996 87
that one can only wonder what their
intended use was! The type numbers
are absent from any of the common
valve catalogs.
Even the bargain price of $1 each, or
$20 a box full, was initially too high
to tempt much interest. But at the end
of the day someone realised their true
worth and took the lot.
Why sell?
A 4-gang tuning capacitor from an ancient TRF receiver. It is made entirely of
brass and the main control shaft rotates on plain bearings.
fairly unlikely but it is an interesting
relic and would make a good display
item.
Another piece of equipment that
had been collecting dust for a few years
is a 1930s Pilot valve tester. It was
bought with the intention of restoring
it and although it is in working condition, the old Pilot has few problems.
First, there are no operating instructions, which is usually the case
with old valve testers. Second, being
a 1930s model, there is no provision
for testing post-war 7-pin and 9-pin
miniature valves, unless one makes up
a few adaptors. And finally, because
the tester is of American manufacture,
it works on 110V and so requires a
step- down transformer for its operation.
While there would be few problems
cleaning up the sockets and switches,
I already have other valve testers,
with operating instructions, and there
seemed little point in keeping this one.
Although the Pilot is usable on early
valves up to octal, perhaps it too would
be better used as display item than as
a working valve tester.
The big swap meet bargain of
bargains was a selection of unique
valves. These valves are so unique
This valve tester was one of a trio of test instru
ments. Presumably the other units were a radio
frequency generator and a volt/ohms/amp meter.
Lack of instructions and 110V operation makes it
fairly unattractive for use as a valve tester.
88 Silicon Chip
Anyone attending a radio swap meet
must wonder why other collectors
want to unload so much of their wares!
If it is so good, why don’t they keep it?
The answer is simple. If a collector
has something he really has no use
for, or he has duplicates of a particular
item, then the answer is to swap, trade
or sell. That way, other things can be
acquired without having to spend
money. It also prevents the accumulation of unwanted junk.
One interesting aspect of a swap
meet is to see what people pay for
the things they want. Most members
of the community would take these
items to the tip and consider them
to be rubbish. Who knows – maybe
they’re right!
To be perfectly honest, after collecting for more than 10 years, I’m starting
to look on some quite collectable receivers as just old radio sets. There is
no reason why I should collect every
While this neat little 1920s receiver looks OK on the outside,
there was quite a lot missing on the inside. It now has a new
owner.
K
alex
The UV People
ETCH TANKS
● Bubble Etch ● Circulating
LIGHT BOXES
● Portuvee 4 ● Portuvee 6
● Dual Level
TRIMMER
● Ideal
PCB DRILL
There’s not much use for old meters such as these now that cheap multimeters
are so readily available. In the distant past, this panel had been used as a volts/
amp test rig.
make and model, nor is there any
reason to have the best of everything.
There is every possibility that over
the next 10 years I will gradually scale
down my collecting activities and reduce the size of my collection, keeping
only the more interesting items. I can’t
take it all with me when I go, can I?
MATERIALS
● PC Board: Riston, Dynachem
● 3M Label/Panel Stock
● Dynamark: Metal, Plastic
✸ AUSTRALIA’S NO.1 STOCKIST ✸
K
alex
40 Wallis Ave, East Ivanhoe 3079.
Phone (03) 9497 3422, Fax (03) 9499 2381
Other throw-outs
I’m getting a bit off the track here.
Let’s get back to clearing out my shed.
One of my other throw-outs was a
1920s 3-valve regenerative receiver in
a neat little cabinet with double doors
at the front, covering the control panel.
I was told it is a Radiola 4 cabinet into
which someone had built the 3-valve
set. Whether that was the case or not it
sold quickly and now has a new owner.
Naturally it had been my intention
to restore the little 3-valver but, as
there are better and more interesting
old regen
erative sets in the shed, I
decided to let this one go.
Accompanying the 3-valve receiver
was a 1926 Brown horn speaker. Horn
speakers are very collectable and although this particular example was a
bit battle scarred it, too, sold quickly.
There are two others in the shed and,
when all is said and done, how many
Brown horns does a bloke need?
Now some of my junk was not
really junk at all but quite nicely restored radio receivers and about half
a dozen mantel radios from the 1940s
and 1950s. Once again, some were
duplicates and I see no need to collect
radios in twos or threes unless one is
into collecting a complete colour range
of a particular model.
● Toyo HiSpeed
Silicon Chip Binders
This Brown horn speaker is one
of the better types in that it has an
aluminium cone instead of the usual
soft iron diaphragm. Its tonal qualities
and sensitivity were better than most.
The restored radios sold very well,
as they were consider
ably cheaper
than those at some of the other sites.
Anything at a fair price will sell. Inflate the price beyond the item’s true
worth and not many buyers will be
forthcoming. I went to the swap meet
to sell, not to bring it all back home
again at the end of the day.
So all things considered, taking a
site at the vintage radio swap meet
proved to be a worthwhile move for
several reasons. It was not only a good
day out whereby I off-loaded some
unwanted equipment but I also met
other collectors whom I would not
SC
have otherwise met.
These beautifully-made binders will
protect your copies of SILICON CHIP.
They are made from a distinctive
2-tone green vinyl & will look great
on your bookshelf.
Price: $A11.95 plus $3 p&p each
(NZ $8 p&p). Send your order to:
Silicon Chip Publications
PO Box 139
Collaroy Beach 2097
Or fax (02) 9979 6503; or ring (02)
9979 5644 & quote your credit card
number.
August 1996 89
PRODUCT SHOWCASE
6/12V Automotive Battery Tester
Just how do you check out your
car’s battery? Unless you try to start
your engine on a very cold morning
you really don’t know if the battery is
up to standard. And when that cold
morning arrives, the battery may fail
when called upon to do its job.
TOROIDAL POWER
TRANSFORMERS
Manufactured in Australia
Comprehensive data available
Harbuch Electronics Pty Ltd
9/40 Leighton Pl. HORNSBY 2077
Ph (02) 476-5854 Fx (02) 476-3231
BassBox®
That’s where the Model 50113 battery tester from Jaycar can be handy.
It can check the battery’s state by
monitoring the voltage with no load
and with a 100-amp load, to simulate
the loading of a typical starter motor
when cranking the battery.
As well as a voltage scale up to
16V, the unit has a colour scale which
grades 6V and 12V batteries in terms
such as “bad”, “weak” and “OK”, as
well as having a scale which corre
lates the voltage under load of a 12V
battery to “cold cranking amps”. This
is a measure of the battery’s ability to
crank the engine when the temperature
is under 5°C. According to the manual
which comes with the tester, a typical
fully charged battery at 5°C has only
40% of the capacity that it possesses
at 25°C.
In practice, the tester is connected
directly across the battery and the
voltage is noted. The red rocker test
button is then pushed for 10 seconds
and the reading on the scale noted.
If the needle is in the red region, the
battery is a dud. Unfortunately, two of
the heavy duty batteries used in our
lab were found wanting in this test;
luckily we were not depending on
them to start a car!
The tester becomes quite warm after
TES sound
level meter
Design low frequency loudspeaker enclosures
fast and accurately with BassBox® software.
Uses both Thiele-Small and Electro-Mechanical
parameters with equal ease. Includes X. Over
2.03 passive crossover design program.
$299.00
Plus $6.00 postage.
Pay by cheque, Bankcard, Mastercard, Visacard.
EARTHQUAKE AUDIO
PH: (02) 9948 3771 FAX: (02) 9948 8040
PO BOX 226 BALGOWLAH NSW 2093
90 Silicon Chip
How quiet is your office or working
environment? For the safety of your
hearing you should not be exposed to
noise levels of more than 85dBA for
long periods at a time. Many factory
environments are much noisier than
this and safety muffs are regarded
as more or less mandatory in such
situations.
Many other working environments
are also quite noisy and are a cause
for concern. Your own home can also
be a hearing hazard, particularly with
such appliances as food mixers and
blenders, vacuum cleaners and even
a load check, as you might expect because it has to dissipate something in
the region of 1000 watts. For this reason, only three such tests are permis
sible in a 5-minute period.
The tester appears to be well made
and is easy to use. It is available from
all Jaycar Electronics stores and re
sellers for $89.95. (Cat QM-1620).
hair-dryers. Many of these can exceed
90dBA and you would be required to
wear hearing protection under normal
conditions of employment. And most
power tools are exceedingly noisy.
Want to check out your own situation? This TES 1350 sound level meter
will do the job. It has a calibrated inbuilt electret microphone and a 4-digit
liquid crystal display which reads in
dB with either “A” or “C” frequency
weighting.
The TES 1350 has two ranges:
Lo, reading from 35-100dB; and Hi,
reading from 65-130dB. The meter
response can be switched to slow
or fast and there is a hold facility to
catch noise peaks. In addition, there
is an inbuilt oscillator which
provides a reference signal for
calibration. This is brought
into play by setting the function slide switch to CAL and
then tweak
ing the adjacent
trimpot to give a reading of
94dB.
The other worthwhile feature of the instrument is that
it has a 3.5mm stereo jack
socket which provides two
output signals for use with
external equipment. One is an
AC output with an impedance
of 600Ω while the other is a
DC logarithmic signal corre
sponding to 10mV/dB. The unit
is powered with a standard 9V
battery with an expected life
of 100 hours (using an alkaline
type).
The TES 1350 is available at
$399 from Altronics, 174 Roe
St, Perth WA 6000. Phone 1
800 999 007.
If you are seeing a blank page here, it is more
than likely that it contained advertising which
is now out of date and the advertiser has
requested that the page be removed to prevent
misunderstandings. Please feel free to visit the
advertiser’s website:
www.emona.com.au/
Spectrum analysers
from Promax
The Promax range of spectrum analysers is available
from Emona Instruments.
There are three instruments in the
range covering the frequency range
up to 1GHz or with an option to cover
up to 1.75GHz, which takes in the
satellite TV IF band.
Key features of the instruments are
automatic selection of the optimum
resolution bandwidth, switchable
input impedance of 50Ω or 75Ω, indication of frequency on a 4½ -digit
display, total dynamic range of 120dB
and built-in calibration.
They offer three vertical axis settings of 2dB/div, 10dB/div and linear
Audiosound home
cinema speakers
detection. Measurement range is
15dBµV to 130dBµV.
For further information, con
tact
Emona Instruments. Phone (02) 519
3933; fax (02) 550 1378.
Audiosound Laboratories has developed three home theatre speaker
system packages which include their
new CE-1 passive equalised centre
channel speaker with double magnetically shielded drivers.
System One is very unobtrusive
and uses the Audiosound space-bass
system incorporating two subwoofers.
The total package comprises seven
loudspeakers in all for under $2000.
August 1996 91
and the CE-1 for the centre. The main
8015s have received an Australian Design Award and this complete package
is well priced at $1890. System Three
is similar to System Two and the same
price but uses the unobtrusive DM-1s
instead of the Piccolos for the rear
channels.
Audiosound are also able to supply complete home theatre packages
including Dolby Pro Logic receivers
and large screen TVs to match.
For further information, contact
Audiosound Laboratories, 148 Pitt
Rd, Curl Curl, NSW 2099. Phone (02)
9938 2068.
Miniature DC motor
can be sterilised
The tiny front and rear speakers come
with wall mounting brackets and can
be colour-matched to order for a highly
unobtrusive total system.
System Two (pictured) uses the
floor-standing 8015s up front, with
their Piccolo system for rear channels
Designed for medical, surgical and
chemical applications, the Maxon
RE 035 40-watt and 2326 6-watt DC
motors can be repeatedly sterilised in
an autoclave. Dismantling is not necessary. They have ironless rotors and
are intended for use in medical hand
tools such as bone saws, drilling and
milling machines, dental and derm
atological equipment, infusion pumps,
therapeutical assistance devices, and
KITS-R-US
PO Box 314 Blackwood SA 5051 Ph 018 806794
TRANSMITTER KITS
•• FMTX1
$49: a simple to build 2.5 watt free running CD level input, FM band runs from 12-24VDC.
FMTX2B $49: the best transmitter on the market, FM-Band XTAL locked on 100MHz. CD level input 3
stage design, very stable up to 30mW RF output.
•• FMTX2A
$49: a universal digital stereo encoder for use on either of our transmitters. XTAL locked.
FMTX5 $99: both FMTX2A & FMTX2B on one PCB.
•connector
FMTX10 $599: a complete FMTX5 built and tested, enclosed in a quality case with plugpack, DIN input
for audio and a 1/2mtr internal antenna, also available in 1U rack mount with balanced cannon
input sockets, dual VU meter and BNC RF $1299. Ideal for cable FM or broadcast transmission over
distances of up to 300 mtrs, i.e. drive-in theatres, sports arenas, football grounds up to 50mW RF out.
FMTX10B $2599: same as rack mount version but also includes dual SCA coder with 67 & 92kHz subcarriers.
•
AUDIO
•soldDIGI-125
Audio Power Amp: this has been the most popular kit of all time with some 24,000 PCBs being
since 1987. Easy to build, small in size, high power, clever design, uses KISS principle. Manufacturing
rights available with full technical support and PCB CAD artwork available to companies for a small royalty.
200 Watt Kit $29, PCB only $4.95.
AEM 35 Watt Single Chip Audio Power Amp $19.95: this is an ideal amp for the beginner to construct;
uses an LM1875 chip and a few parts on a 1 inch square PCB.
Low Distortion Balanced Line Audio Oscillator Kit $69: designed to pump out line up tone around studio
complexes at 400Hz or any other audio frequency you wish to us. Maximum output +21dBm.
MONO Audio DA Amp Kit, 15 splits: $69.
Universal BALUN Balanced Line Converter Kit $69: converts what you have to what you want, unbalanced
to balanced or vice versa. Adjustable gain. Stereo.
•
•
••
COMPUTERS
•to Max
I/O Card for PCs Kit $169: originally published in Silicon Chip, this is a real low cost way to interface
the outside world from your PC, 7 relays, 8 TTL inputs, ADC & DAC, stepper motor drive/open collector
1 amp outputs. Sample software in basic supplied on disk.
•onlyIBM3 chips
PC 8255 24 Line I/O Card Kit $69, PCB $39: described in ETI, this board is easy to construct with
and a double sided plated through hole PCB. Any of the 24 lines can be used as an input or
output. Good value.
•• Professional
19" Rack Mount PC Case: $999.
All-In-One 486SLC-33 CPU Board $799: includes dual serial, games, printer floppy & IDE hard disk drive
interface, up to 4Mb RAM 1/2 size card.
•PC104
PC104 486SLC CPU Board with 2Mb RAM included: 2 serial, printer, floppy & IDE hard disk $999; VGA
card $399.
KIT WARRANTY – CHECK THIS OUT!!!
If your kit does not work, provided good workmanship has been applied in assembly and all original parts
have been correctly assembled, we will repair your kit FREE if returned within 14 days of purchase. Your
only cost is postage both ways. Now, that’s a WARRANTY!
KITS-R-US sell the entire range of designs by Graham Dicker. The designer has not extended his agreement
with the previous distributor, PC Computers, in Adelaide. All products can be purchased with Visa/Bankcard
by phone and shipped overnight via Australia EXPRESS POST for $6.80 per order. You can speak to the
designer Mon-Fri direct from 6-7pm or place orders 24 hours a day on: PH 018 80 6794; FAX 08 270 3175.
92 Silicon Chip
analytical and dialysis equipment.
Both can be vapour sterilised to
135°C and are pressure insensitive
to 3.6 bar in 100% relative humiity.
The 40-watt RE035 has a diameter
of 35mm, is 71mm long and has an
efficiency of 82%. The smaller 6-watt
model is 26mm in diameter, 44mm
long and has an efficiency of 75%.
Both are easy to speed control and
have a speed range from 0-11,000
rpm.
For further information, contact M.
Rutty & Co, 4 Beaumont Rd, Mt Kuringai, NSW 2080. Phone (02) 457 2222.
Static RAM has on-board battery
A new range
of modules from
Benchm arq incorporate a static
RAM with onboard battery,
real-time clock
and CPU supervisor and directly replaces
industry standard 28-pin static
RAMs. In effect,
the Benchmarq bq4830 provides non-volatile static
RAM by combining an internal lithium battery with
a 32K x 8 CMOS SRAM, a quartz crystal clock and a
power-fail chip. It provides 10-year minimum data
retention and unlimited write cycles.
The bp4832 provides full CPU supervision plus the
features of the bp4830 in a 32-pin package. It provides
a watchdog timer, power-on reset, alarm/periodic
interrupt, power-fail and low battery warning. The
bp4842 has all the features of the bp4832 but is a
128K x 8 SRAM as well.
For further details, contact Reptechnic, 3/36 Bydown St, Neutral Bay, NSW 2089. Phone (02) 9953
9844; fax (02) 9953 9683.
ASK SILICON CHIP
Got a technical problem? Can’t understand a piece of jargon or some technical principle? Drop us a line
and we’ll answer your question. Write to: Ask Silicon Chip, PO Box 139, Collaroy Beach, NSW 2097.
Slo-blo fuses
not desirable
I am keen to buld the 175W amplifier
module described in the April 1996
issue of SILICON CHIP. However, a
mate of mine suggested that it should
be fitted with slow-blow fuses in the
supply lines to avoid the nuisance of
fuses blowing at inconvenient times.
I notice that you have not specified a
particular type of fuse in the parts list,
so should I take my mate’s advice. (B.
L., Campsie, NSW).
• While the transistors in this module
are quite rugged, there is only one line
of protection in the circuit – the fuses.
If the current gets to the point where
the fuses should blow, you don’t want
any delay, otherwise the transistors
will blow before the fuses. In fact, as
a general rule, fuses which protect
semiconductor circuits should alway
be standard fast-blow types.
Slow-blow fuses are normally specified only where the component to be
protected can easily withstand large
current overloads and is subjected
to them in normal operation. In such
cases, a normal fuse would be subject
to nuisance blowing. A good example
of this is with large toroidal power
Speed control for
cassette recorders
I am teaching myself to play the
mandolin and find it helpful trying
to play along with a tape cassette
and recordings. The problem is
that many recordings are not in an
exact key; ie maybe a quarter of a
note sharp or flat of a true key (such
as half way between A and A#). I
therefore need a system to slow or
speed the cassette motor up by only
a small frequency shift.
If there is no easy way, perhaps
you could publish some details on
the theory of tape motor control. (C.
F., Cringila, NSW).
[dot]In most tape recorders the
transformers which draw a large initial
surge current. See our article on fuses
for these transformers in the March
1995 issue of SILICON CHIP.
Having poo-poohed slow blow fuses, we are planning a PA version of this
175W amplifier and it will incorporate
short circuit protection.
Colour TV
pattern generator
I have just completed building the
Colour TV Pattern Generator and have
run into some problems.
First off, I’m operating the unit via
a variable-voltage battery eliminator
plugpack. This unit supplies DC at 3,
4.5, 6, 7.5, 9 and 12V at 300mA. I hope
this has nothing to do with the fault
I’m about to describe. It took me the
better part of two days to construct the
kit and what seemed like 4 or 5 hours
wiring in those bridging connections
(if I never see another bridge again,
it’ll be too soon).
I took every precaution to ensure
that all the components were wired
in correctly, especially the electrolytic
capacitors, diodes and such where
polarity is important. When I finally
got to test the system, I found I could
capstan motor is a DC type with
some sort of tachometric speed
control via an extra pair of wind
ings. In some cases, if you have
access to the circuit dia
gram, it
may be possible to use a trimpot to
vary the tacho signal, and thereby
the speed of the motor.
Failing that, you could decide
to separately power the capstan
motor via an external speed control
circuit. This might not have the
same degree of speed regulation
but at least you would be able to
control it yourself.
As a starting point, you might
consider the speed control published on page 15 of the August
1992 issue.
get the Check, Hatch, Dot and White
Raster OK but the Red Raster and Colour Bars/Greyscale would not operate
properly. The Red Raster was erratic
and the Colour Bars weren’t there at
all; neither was the Greyscale.
I said the first four operated OK,
although not absolutely perfectly
because a descending horizontal line
would appear across the screen and
where it appeared, the pattern would
appear to bend behind it. Are there any
component value changes I should be
aware of (resistors, capacitors) or any
other components that might have
been supplied that were wrong for
this kit?
My purpose in constructing this
kit is not for TV servicing work but
to insert colour bars at the head of,
between items and on the tail-end
of videocassettes recorded off-air or
from a video camera. (N. F., Stockton,
NSW).
• It is important that the plugpack
is rated at 500mA to prevent the regulator (12V) from dropping out and
producing mains hum in the supply.
A larger value electrolytic than the
470uF capacitor at the input to REG1
will also help.
Problems with the greyscale and
colour bars can be caused by IC14 not
oscillating or IC15 not counting. Also
check that IC13b has a high Q output
at pin 9 when the bars are selected by
switch S2a.
Cat deterrent
not humane
May I suggest a project which
would, I believe, be of great practical
use. There are reports that bells around
cats’ necks do little or nothing to prevent the slaughter of native animals
and birds by family pets, put out, as
is usual, at night.
My idea would be for a small
electronic unit to be worn by the cat,
continuously emitting a pulsed tone
at a supersonic frequency but within
the audio frequency range of small
mammals and birds.
August 1996 93
Questions on digital
signal generator
Would you please supply me
with some information on the
Digital Signal Generator kit published in the July 1990 issue of
SILICON CHIP. I don’t have access
to the article and am totally ignorant on its internal workings. I am
assuming that in its 10Hz-1000Hz
range it displays frequency in 1Hz
increments. I would like to know
if it actually generates frequency
in between these 1Hz increments
such as 30.25Hz or 30.2Hz etc.
I need an affordable signal generator that can do this and at the same
time have solid amplitude stability.
I will be using it in establishing the
Thiele-Small loudspeaker parameters and driver/box design. (S. W.,
Modbury North, SA).
• The frequency output of the
Digital Signal Generator is continuously adjustable over its entire range from 0.1Hz to 500kHz
but its display only has 4-digit
resolution. This has two ramifications. First, on the 10Hz to
1000Hz range, it should display
frequencies between 9.9Hz and
999.9Hz although the actual range
will depend on the initial setup of
the instrument.
I don’t know what the frequency
would be but perhaps someone has
done or could do a few experiments,
using the silent dog whistle as a starting point. A frequency that is inaudible
to the cat but audible to its prey would
be the ideal.
As the range would need to be only
a few feet, the output power should be
low and could, I hope, be supplied by
one of the larger watch batteries. The
unit could then be attached to the collar in place of the conventional bell. It
would, of course, spell the end of the
cat’s days as a mouser! (K. F., Albion
Park Rail, NSW).
• We don’t think a warning device
to prevent cats from catching birds
would work. While ultrasonics can be
used with great effect to deter cats and
dogs from entering an area, we have
no evidence that ultrasonics affects
birds at all.
We also think that fitting a cat with
94 Silicon Chip
Second, the accuracy of the frequency display is ±2% + 2 digits
which means that a frequency read
out of 512.5Hz is not an absolute
figure; the actual frequency could
be anywhere between 500Hz and
523Hz.
And while the frequency can be
varied continuously, it is doubtful
whether the mechanical resolution
of the potentiometers would let you
reliably and repeatedly set a particular frequency of, say, 512.5Hz.
And even if you did manage this
feat, it is doubtful whether the
frequency stability of the instrument would be able to hold that
particular figure for any length of
time, say 15 minutes.
In any case, you don’t need great
frequency accuracy if you are working out Thiele-Small parameters.
A figure of ±2% would be quite
adequate. When measuring speaker
resonances, it is not possible to determine them with great accuracy
since they vary with the amplitude
of the drive signal, the temperature
and with the age of the speaker. The
digital signal generator is certainly
adequate for this task.
If you really needed the degree
of frequency accuracy and stability
you describe, you would have to
pay many thousands of dollars.
such a device would be cruel. In our
opinion, cats should not be put out at
night, otherwise they will definitely
slaughter native animals and birds.
Of course, cats kill birds, lizards, etc
at any time of the day and so people
should probably think twice about
having a cat in the first place.
Digital display for
Geiger counter
Having recently completed the
Geiger Counter published in your
October 1995 issue, I wonder if others feel as I do that some analog or
digital measuring system registering
the counts/second would improve its
versatility and interest? I suggest that,
using the chart on page 16, it would
be best calibrated in rads/hour. (N. A.,
Hamilton, Qld).
• We plan to publish a modified circuit from one of our contributors in
the “Circuit Notebook” pages of the
coming September issue.
How to solder mask
PC boards
Some time ago I purchased a software package, namely EASYPC, for the
fabrication of PC boards. It has proved
to be tremendous but in the program
there is provision for the laying down
of a solder-resist mask.
Could you please advise if there is a
photo sensitive, proprietary product,
that allows one to lay down the solder
resist mask, in the same manner as the
tracks are laid down? The drilling of
such circuit boards is, to say the least,
irksome and whilst there is a drilling
program within EASYPC, it requires
a numerically controlled drill to implement it.
Would it be possible to run such a
project? I’m sure that there are others
such as I, who like to design and
manufacture their own boards. (N. B.,
Townsville, Qld).
• As far as we know, solder masks
are applied to finished PC boards by a
silk-screen process. Component overlays are printed by the same process.
We don’t know of any photo-sensitive
product. However, perhaps one of our
readers may know of a product.
Tone controls for a
guitar amplifier
I was recently given a 300W amplifier kit and I thought I might make
a guitar amplifier out of it. I am now
looking for a tone control circuit
consisting of treble, middle, bass,
gain master volume and many other
special items before I start building
the amplifier.
I am also looking for circuit diagrams for a distortion foot pedal. Other
circuit diagrams for foot pedals such
as the “Flanger” would also be greatly
appreciated. Could you give me some
advice about a suitable loudspeaker as
well? (X. Z., Punchbowl, NSW).
• We have published two projects
which are relevant to your request: a
five band equaliser in December 1995
and a digital effects unit in February
1995. In addition, we published a
three-band tone control (circuit only)
in the Circuit Notebook pages of the
February 1991 issue. We can supply
any of these back issues for $7 each,
SC
including postage.
MARKET CENTRE
Cash in your surplus gear. Advertise it here in Silicon Chip.
FOR SALE
CLASSIFIED ADVERTISING RATES
Advertising rates for this page: Classified ads: $10.00 for up to 12 words plus 50
cents for each additional word. Display ads (casual rate): $25 per column centimetre (Max. 10cm). Closing date: five weeks prior to month of sale.
To run your classified ad, print it clearly in the space below or on a separate
sheet of paper, fill out the form & send it with your cheque or credit card details
to: Silicon Chip Classifieds, PO Box 139, Collaroy, NSW 2097. Or fax the details
to (02) 979 6503.
_____________ _____________ _____________ _____________ _____________
_____________ _____________ _____________ _____________ _____________
_____________ _____________ _____________ _____________ _____________
_____________ _____________ _____________ _____________ _____________
_____________ _____________ _____________ _____________ _____________
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_____________ _____________ _____________ _____________ _____________
_____________ _____________ _____________ _____________ _____________
Enclosed is my cheque/money order for $__________ or please debit my
❏ Bankcard ❏ Visa Card ❏ Master Card
Card No.
✂
_____________ _____________ _____________ _____________ _____________
SATELLITE DISHES: international
reception of Intelsat, Panamsat, Gori
zont,Rimsat. Warehouse Sale – 4.6m
dish & pole $1499; LNB $50; Feed $75.
All accessories available. Videosat, 2/28
Salisbury Rd, Hornsby. Phone (02) 482
3100 8.30-5.00 M-F.
WEB Search on ‘Dontronics’ for 18,
24, 28, 40 and 48-pin ZIF sockets.
MicroChip PIC items: CPUs, Basic
Compilers, Interpreters, Programmers
from $20, Real Time Clock, A-D. Ring or
fax for Free Promo Disk. 29 Ellesmere
Crescent, Tullamarine 3043. Phone (03)
9338 6286. Fax (03) 9338 2935.
RAIN BRAIN 8 STATION SPRINKLER
KIT: Ultra reliable & versatile Hi Q kit.
Rain switch & LED B/L Free!!! (SC Jan.
1996). Mantis Micro Products, 38 Garnet St, Niddrie, 3042 P/F/A (03) 9337
1917 mantismp<at>c031.aone.net.au
MINILOG KIT from MicroZed $25 incl.
S/T, programs on disk, all parts except
BS2.
C COMPILERS: Dunfield compilers are
now even better value. Everything you
need to develop C and ASM software
for 68HC08, 6809, 68HC11, 68HC16,
8051/2, 8080/85, 8086 or 8096: $140.00
each. Macro Cross Assemblers for these
CPUs + 6800/01/03/05 and 6502: $140
for the set. Debug monitors: $70 for 6
RCS RADIO PTY LTD
Signature__________________________ Card expiry date______/______
Name ______________________________________________________
Street ______________________________________________________
Suburb/town ___________________________ Postcode______________
RCS Radio Pty Ltd is the only company that manufactures and sells every
PC board and front panel published
in SILICON CHIP, ETI and EA.
RCS Radio Pty Ltd,
651 Forest Rd, Bexley 2207.
Phone (02) 587 3491
August 1996 95
MicroZed Computers
PO Box 634, ARMIDALE 2350 (296 Cook’s Rd)
Ph (067) 722 777 – may time out to Mobile 014 036 775
Fax (067) 728 987 (Credit Cards OK)
Specialising in easy-to-get-going hard/software kits.
Send 2 x 45c stamps for information package
Microchip
Programmers, Simulators and PIC chips
NEW Micro
68HC11 F1 boards and now 80535 (up spec 8051)
Extra I/O and peripheral plug-ins too
Micro Engineering Labs
BASIC compiler for PIC and PIC programmers
Advertising Index
Av-Comm.....................................71
BASIC Stamp I and II
Macintosh patch now available
Car Projects Book....................OBC
T
Scott Edwards Electronics
Dick Smith Electronics........... 10-13
OPTO 22
Emona.........................................91
ngamebobs
hiAustralian
kits
Accessories for Stamp and second source for Stamp 1
Earthquake Audio........................90
Optically isolated drivers for AC & DC
Freedman....................................81
MEMORY * DRIVES * MODEMS
SPECIAL! (ExTax)
1Mbx9 – 70ns
$25
30-pin Simms
CPUs. All compilers, XASMs and monitors: $400. 8051/52 or 80C320 simulator
(fast): $70. NEW: Disassemblers for
12 CPUs only $75. Demo disk: FREE.
All prices + $5 p&p. GRANTRONICS
PTY LTD, PO Box 275, Wentworthville
2145. Ph/Fax (02) 631 1236 or Internet:
lgrant<at>mpx.com.au.
EPROM PROGRAMMER FOR SALE:
GTEK system. Programs up to 8
EPROMs at one time. $300. Call Sandro
on (02) 757 2543.
MICRO ENGINEERING LABS PBASIC
Compiler $120 from MicroZed + $5 post.
Put Stamp programs into raw PIC chips.
KITS KITS KITS: PC printer port Relay
Board with DOS/WIN drivers $68.50. DC
Speed Controller $33.15, 110db Piezo
Screamer $19.90. IR Toggle Switch
$18.40, CCD cameras $185.00. FM
Trans
mitters, Amplifiers, Power Supplies, Microcontroller kits and more.
FREE catalog available. Ozitronics,
24 Ballandr y Crescent, Greens
borough 3088. (03) 9434 3806.
ozitronics<at>c031.aone.net.au
http://www.hk.super.net/-diykit/oz.html
MICROCRAFT PRESENTS: Dunfield
(DDS) products are now available exstock at a new low price; please ask for
our catalogue. Micro C, the affordable
SIMMS
(Parity/No Parity)
4Mb 30 PIN-70
$71
$90
4Mb 72 PIN-70
$75
$53
8Mb 72 PIN-70
$133 $100
16Mb 72 PIN-70 $230 $192
32Mb 72 PIN-70 $456 $378
EDO SIMMS
8Mb (1Mbx32) – 60ns $118
16Mb (2Mbx32) – 60ns $210
MAC MEMORY
8Mb P’BOOK 190 $240
VIDEO MEMORY
256K x 16 70ns (SOJ) $17
256K x 16 70ns (ZIP) $48
LASER PRINTER MEMORY
2Mb UPGRADE
$140
CO-PROCESSORS
80387SX/DX to 40MHz
$100
COMPAQ
8Mb CONTURA AERO
$240
All other models available $Call
TOSHIBA PORTEGE/SATELLITE
8Mb / 16Mb EDO $294 / $550
All other models available $Call
IDE DRIVES: SEAGATE/CONNER
1080Mb EIDE 10.5ms 3yr $283
1620Mb EIDE 14ms 3yr $360
2113Mb EIDE 10.5ms 3yr $384
MODEMS: BANKSIA / SPIRIT
28,800 BANKSIA V.34
$360*
28,800 SPIRIT V.34/V.FC $350*
*Plus 14% sales tax on modems
Ex Tax Pricing – Delivery $8. Pricing as at 26/6/96. Phone for latest.
Sales Tax On Modems 14%. Everything Else 22%.
Credit Cards Welcome. We Also Buy And Trade-In Memory.
PELHAM
Memory Pty Ltd
Suite 6, 2 Hillcrest Rd,
Ph: (02) 9980 6988
Pennant Hills, 2120.
Fax: (02) 9980 6991
Email: pelham1<at>ozemail.com.au
Instant PCBs................................96
Jaycar ................................... 45-52
Kits-R-US.....................................92
Latrobe University........................27
Macservice............................ 28-29
MicroZed Computers...................96
Model Railway Projects Book......18
Oatley Electronics.....................3,85
“C” compiler for embedded applications.
Versions for 8051/52, 8086, 8096,
68HC08, 6809, 68HC11 or 68HC16
$139.95 each + $3 p&h • Now on special is the SDK, a package of ALL the
DDS “C” compilers for $399 + $6 p&h •
EMILY52 is a PC based 8051/52 high
speed simulator $69.95 + $3 p&h • DDS
demo disks $7 + $3 p&h • VHS VIDEO
from the USA (PAL) “CNC X-Y-Z using
car alternators” (uses car alternators as
cheap power stepper motors!) $49.95
+ $6 p&h (includes diagrams) • Device
programming EPROMs/PALs etc from
$1.50 • Fixed price electronic design and
PCB layout • Credit cards accepted • All
goods sent certified mail • Call Bob for
more details. MICROCRAFT, PO Box
514, Concord NSW 2137. Phone (02)
744 5440 or fax (02) 744 9280.
MicroZed HAVE range of PIC chips.
OTP and /JW versions available. PIC
16C84 /04 one off price $9.76 incl. S/T.
Microprocessors For Silicon Chip Circuits
We have stocks of the 68HC705-C8P pre-programmed microprocessor ICs for the Digital Effects Unit
(February 1995) and the Remote Controlled Stereo Preamplifier (Sept.-Oct. 1993). Also available is the
pre-programmed Z86E08 microprocessor for the Railpower Mk.2.
68HC705-C8P – $45 ea; Z86E08 $18 ea. Prices include p&p.
Payment by cheque, money order or credit card to: Silicon Chip Publications, PO Box 139, Collaroy, NSW
2097. Phone (02) 9979 5644; Fax (02) 9979 6503.
96 Silicon Chip
Harbuch Electronics....................90
Pelham........................................96
RCS Radio ..................................95
Rod Irving Electronics .......... 59-63
Scan Audio..................................75
Silicon Chip Binders....................39
Silicon Chip Bookshop.................53
Silicon Chip Software..................37
Tortech.........................................27
Tektronix....................................IFC
Zoom.........................................IBC
_________________________________
PC Boards
Printed circuit boards for SILICON
CHIP projects are made by:
• RCS Radio Pty Ltd, 651 Forest
Rd, Bexley, NSW 2207. Phone (02)
587 3491.
• Marday Services, PO Box 19-189,
Avondale, Auckland, NZ. Phone (09)
828 5730.
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