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This amplifier is capable of delivering over 500 watts into 4Ω or around 280
watts into an 8Ω load. The large heatsink is mandatory and needs to be fancooled if it is to withstand the rigours of operating under maximum dissipation
conditions. We envisage it as being used in high-end stereo systems and for
musical instrument and PA work.
500W
of audio power
24 Silicon Chip
W
N
Pt.1: By LEO SIMPSON & BOB FLYNN
O MATTER WHICH WAY you
look at it, this is a big power
amplifier. It’s physically big,
it needs a big power supply and a
big fan-cooled heatsink and it delivers lots of power. A pair of these
amplifiers would be the basis of a
magnificent stereo system for the
home, especially if you have a large
listening room.
Perhaps you might think that a
500 watt per channel stereo system
would be too much. The answer to
that depends on what sort of music
you like listening to and how efficient
your loudspeakers are. If you like rock
music with its fairly limited dynamic
range (ie, loud all the time), then a
1000 watt system would be going
over the top. But if you listen to a
lot of classical piano music and your
speakers are of only average efficiency, then 500 watts per channel might
not be enough!
One of the authors of this article
has a large piano in his (large) loungeroom and often has the opportunity
(every day) to compare the real piano
with CDs played through the Studio
200 power amplifier published in the
February 1988 issue of SILICON CHIP.
That amplifier has a music power
output of 120 watts per channel into
8Ω and 190 watts per channel into
4Ω. In a straight comparison for absolute loudness and dynamic range,
the real live piano, played by an accomplished pianist, wins every time.
We’re not talking about ridiculously loud music here – just a
piano competently played. What is
not commonly realised is that the
piano is probably the most difficult
musical instrument to accurately
Do you want a big power amplifier for musical instrument
or PA use? Something with real grunt? Well here it is, the
biggest power amplifier ever described in an Australian
magazine and probably the biggest published anywhere in
recent years. It delivers 500 watts RMS into a 4Ω load and
278 watts into an 8Ω load.
August 1997 25
26 Silicon Chip
Fig.1: the circuit uses 12 output
transistors in a complementary
symmetry arrangement, driven by
an MJL21193/4 pair; ie, the same as
the output transistors. Short circuit
current limiting is provided by Q24
& Q25. The supply rails are ±80V so
we have had to specify high voltage
transistors for the input differential
pair, Q1 & Q2.
record and reproduce because of its
huge dynamic range – even when it’s
not being played particularly loudly,
most amplifiers and loudspeakers are
not up to the task. But a pair of these
new power amplifiers and large loudspeakers to match would certainly
cope with any CD of classical piano!
Without getting too much ahead of
ourselves, this new amplifier design
produces only about 5dB more power
than the 1988 design so the difference
in absolute loudness won’t be huge.
On the other hand, it will be noticeably louder and will be far less likely to
be over-driven than the older design.
Background to the design
It’s been a long time coming, this
amplifier. It was first mooted more
than 12 months ago in 1996 and we
have made several false starts since,
only to come to a stop as component
availability or suitability stopped us
from proceeding further.
Also along the way we produced
a full bridge design, effectively two
power amplifiers on the one PC board
which drive the single loudspeaker
in anti-phase. The driving voltages
from the two amplifiers add and so
the power delivered is the sum of
the power outputs from the two amplifiers. The advantage of the bridge
design is that the amplifier supply
voltages can be substantially less than
the equivalent large single-ended
amplifier.
The lower supply voltages mean
that the electrolytic capacitors in the
power supply are less expensive and
the transistors used throughout the
amplifier can have a lower voltage
rating. In practice, it was the rarity
of suitable high voltage high current
driver transistors that pushed us
along this line of development.
However, the resulting bridge amplifier proved to be not as efficient as
a single-ended design and with the
heatsink available to us at the time,
Fig.2: these are the load lines for 4Ω and 8Ω operation. The straight lines are for
resistive loads while the arched lines are for reactive 4Ω (2.83Ω + j2.83Ω) and
8Ω (5.6Ω + j5.6Ω) loads. The concave lines show the 1200W power hyperbola
(dotted) and the one-second SOAR curve for six MJL21193/4 power transistors.
As you can see, the reactive 4Ω load comes quite close to the one-second SOAR
curve. That is why a total of 12 output power transistors is required.
it proved impossible to cool it effec
tively, even with two fans!
After running up that blind alley,
we went back to the drawing board.
This time we were successful, with
a bigger heatsink, fan cooling and a
thermal cutout. And instead of using
conventional driver transistors, we
used power output transistors in the
driver stages. The power transistors
specified have the advantage of being much more rugged and with a
minimum gain-bandwidth product
of 4MHz, their high frequency performance is just as good as many driver
transistors such as the commonly
used Motorola MJE340/350 pairs.
The result of all the development
Specifications
Output power....................................278 watts into 8Ω; 500 watts into 4Ω
Music power.....................................315 watts into 8Ω; 590 watts into 4Ω
Frequency response ........................-0.3dB at 20Hz and 20kHz (see Fig.8)
Input sensitivity.................................1.43V RMS (for full power into 8Ω)
Harmonic distortion..........................typically less than .01%
Signal-to-noise ratio............................... 117dB unweighted (20Hz - 20kHz);
122dB A-weighted
Damping factor.................................>170 at 100Hz & 1kHz; >75 at 10kHz
Stability.............................................unconditional
August 1997 27
AUDIO PRECISION SCTHD-W THD+N(%) vs measured
10
LEVEL(W)
19 JUN 97 22:07:52
1
0.1
0.010
0.001
10
100
800
Fig.3: THD (total harmonic distortion plus residual noise) versus power at 1kHz
into a 4Ω load.
AUDIO PRECISION SCTHD-W THD+N(%) vs measured
10
LEVEL(W)
19 JUN 97 22:09:02
1
0.1
0.010
0.001
10
100
800
Fig.4: THD (total harmonic distortion plus residual noise) versus power at 1kHz
into an 8Ω load.
work is an amplifier capable of delivering 500 watts into a 4Ω load at
.04% harmonic distortion and 278
watts into an 8Ω load at less than
.009% harmonic distortion. Using
the IF Music Power test conditions,
the power output is 590 watts into
4Ω and 314 watts into 8Ω.
Big power like this does not come
in small packages. The amplifier uses
28 Silicon Chip
14 power transistors in all, from the
Motorola MEL21193/94 series. These
plastic power transistors are rated at
250 volts, 16 amps (30 amps peak)
and 200 watts and have been featured
in previous amplifier designs in the
April 1996 and March 1997 issues of
SILICON CHIP.
As indicated above, two of the
power transistors are used as drivers
while the other twelve are used in
the output stage. All are mounted on
a large single sided heatsink. The PC
board measures 362 x 99mm.
This month we are presenting
just the PC board module itself but
because of its sheer size and power
output we strongly recommend that
readers do not “do their own thing”
and install the module with just any
old power supply components and in
just any old chassis. So next month
we will present the full details of
mounting the PC module in a chassis
with a big power supply, fan cooling,
the overload protection module presented in April 1997 and so on. By
the way, we will be presenting it as a
rack mounting mono amplifier only;
if you want that magnificent stereo
setup mentioned above, you would
need two of these mono amplifiers.
Performance
The main performance parameters
are summarised in the accompanying
specifications panel and also demonstrated in a number of graphs. These
indicate that just because a power amplifier delivers a lot of power it does
not mean that it cannot deliver high
performance as well. This amplifier is
very quiet (-122dB A-weighted with
respect to full power into 8Ω) and has
low distortion, typically around .01%
or less. In fact, the amplifier is quieter
than any CD player on the market.
Note that there is not a lot of difference between the music power output
and the continuous power output of
this amplifier; ie, 500W continuous
versus 590W music power. This
amounts to a “dynamic headroom”
figure of 0.7dB for 4Ω loads. This
is a reflection of the fact that the
power supply is very well regulated
– a consequence of using an 800VA
transformer and a filter capacitor
bank of 80,000µF in total. While this
may seem extravagant, cutting back
on the power supply parameters does
prejudice the performance.
Note also that our power figures are
quoted for a mains supply voltage of
240VAC. Typically, the mains supply
is higher than this and so the maximum “unclipped” power output will
be somewhat higher again.
Bipolars vs. Mosfets
In line with our philosophy of
generally not using Mosfets in audio
amplifiers, we have used bipolar tran-
sistors in the output stages. Bipolar
transistors have the advantage of
requiring a lower quiescent current
(to avoid crossover distortion) and for
a given supply voltage they deliver
more power than an equivalent design using Mosfets. Bipolars are also
generally cheaper than equivalent
complementary Mosfets (ie, N-channel and P-channel pairs).
Furthermore, as a result of our
recent testing of this amplifier under conditions of maximum power
dissipation, we are convinced that
a Mosfet amplifier of this power
rating would require considerably
larger fan-cooled heatsinks if it was
to be able to deliver its rated power on a continuous basis. Mosfet
amplifiers are reputed to be almost
“unburstable” because if they become
overheated, they tend to shut down.
While this is an advantage under
overload conditions, this characteristic is a drawback when you want
the amplifier to deliver lots of power
on a continuous basis. As a Mosfet
amplifier gets hotter, it delivers less
power. If it gets very hot, it throttles
right back.
By contrast, if a bipolar design becomes very hot, it still keeps on delivering the goods and the heatsink must
prevent the output transistors from
becoming overheated otherwise they
will be destroyed. Overall though, a
bipolar design is more efficient and
requires less heatsinking.
AUDIO PRECISION SCTHD-HZ THD+N(%) vs FREQ(Hz)
5
19 JUN 97 22:46:21
1
0.1
0.010
0.001
20
100
1k
10k
20k
Fig.5: THD versus frequency at 250W RMS into a 4Ω load.
AUDIO PRECISION SCTHD-HZ THD+N(%) vs FREQ(Hz)
5
19 JUN 97 22:44:35
1
0.1
Circuit details
The full circuit diagram is shown
in Fig.1. Aside from the large number
of output transistors, the circuit is
almost identical in configuration to
the lower power designs featured in
April 1996 and March 1997. It also
incorporates the same short-circuit
overload protection circuit as in the
March 1997 design.
For the benefit of those readers who
have not seen the previous articles
and for the sake of completeness we
shall go through the circuit description in detail.
Note that the supply rails are ±80V
or a nominal 160V in total, under
no signal conditions. This very high
voltage has required us to specify
more rugged transistors than have
been required in the past. This is particularly the case for the driver transistors, as already mentioned, and for
the input transistor pair, Q1 & Q2. In
0.010
0.001
20
100
1k
10k
20k
Fig.6: THD versus frequency at 150W RMS into an 8Ω load.
the latter case, we have specified two
2N5401s rather than the BC556s we
have used in the past. The 2N5401s
have a collector voltage rating of 150
volts versus 80 volts for the BC556.
The input signal is coupled via a
2.2µF capacitor and 1.2kΩ resistor to
the differential pair of transistors Q1
& Q2. Q3 is a constant current source
which sets the current though the
differential pair. The current through
Q3 is set by diodes D1 & D2 and this
sets the voltage across Q3’s 120Ω
emitter resistor to 0.85V. This sets the
current though Q3 to 7mA and so this
is shared by Q1 & Q2 at 3.5mA each.
Q3 is included instead of a common
emitter “tail” for Q1 & Q2 because it
renders the amplifier less sensitive to
variations in the power supply rails.
This is known as PSRR (power supply
rejection ratio) and all good amplifier
August 1997 29
AUDIO PRECISION SCFRQRES AMPL(dBr) vs FREQ(Hz)
5.0000
19 JUN 97 22:40:55
4.0000
3.0000
2.0000
1.0000
0.0
-1.000
-2.000
-3.000
-4.000
-5.000
20
100
1k
10k
20k
Fig.7. frequency response at 20W into a 4Ω load.
AUDIO PRECISION SCFRQRES AMPL(dBr) vs FREQ(Hz)
5.0000
19 JUN 97 22:42:24
4.0000
3.0000
2.0000
1.0000
0.0
-1.000
-2.000
-3.000
-4.000
-5.000
20
100
1k
10k
20k
Fig.8: frequency response at 10W into an 8Ω load.
designs, including op amps, feature
a very high PSRR.
Current mirror
The collector loads of Q1 & Q2 are
provided by Q4 & Q5 which operate
as a “current mirror”. While it is a
little hard to visualise just how a
“current mirror” works, it is easier
if you think of Q5 acting as a sharp
30 Silicon Chip
cutoff diode, providing a voltage at
the base of Q4 which is equal to the
base-emitter voltage drop of Q5 (about
0.6V) plus the voltage drop across its
220Ω emitter resistor.
What happens is that if Q2 tends to
draw more than its share of emitter
current from Q3, the voltage at the
base of Q4 tends to increase and so
Q4’s collector current tends to rise
also. This forces Q1 to pull a bit
more current and stop Q2 from taking
more that its fair share. We say that
Q4 “mirrors” Q5 and so Q1 “sees”
a collector load which is a higher
impedance than would otherwise
be the case. The result is increased
gain and improved linearity from the
differential input stage.
As a matter of interest, current mirror stages are very commonly used in
op amp ICs, partly because they are
easy to design in and partly because
of their enhanced performance.
The signal from the collector of Q1
drives a cascode stage comprising
transistors Q7 & Q8, together with
the constant current load transistor
Q6 (top of the circuit). The cascode
stage is another circuit which is a
little hard to visualise but if you break
it into sections, it is easier.
Note that Q8 has a 3.3V zener diode
ZD1 to hold its base voltage constant
and so Q8 acts like an emitter follower
to provide a constant collector voltage
to Q7. This eliminates any gain variations (non-linearities) which would
otherwise occur if Q7’s collector
voltage was free to vary.
The varying current drawn by
Q7 becomes the input signal to the
emitter of Q8 which is effectively
operating as a “grounded base”
stage. Q8 converts the varying signal
current at its emitter into a varying
signal voltage at its collector. The
combined effect of operating such a
cascode stage is improved linearity
and bandwidth compared with a
single common emitter stage.
A 100pF capacitor from the collector of Q8 to the base of Q7 rolls off
the open loop gain of the amplifier
to ensure a good margin of stability;
ie, to eliminate the possibility of the
amplifier oscillating supersonically.
The output from the cascode stage
is coupled to the driver transistors,
Q10 & Q11. As mentioned previously,
these are MJL21193/94 power transistors, the same as in the output stage.
Note that the signals to the bases of
Q10 & Q11 are identical, apart from
the DC offset provided by Q9.
Vbe multiplier
In setting the DC offset between
Q10 & Q11, Q9 is actually setting the
quiescent current in the output stages.
It provides a forward bias of about
2.3V or so between the bases of Q10
& Q11 so that they are always slight-
ly turned on, regardless of whether
signal is present or not; that is why it
is referred to as “quiescent” current.
Q9 acts as a “Vbe multiplier”,
multiplying the voltage between its
base and emitter by the ratio of total
resistance between its collector and
emitter to the resistance between its
base and emitter.
In practice, trimpot VR2 is adjusted not to give a particular voltage
between the collector and emitter of
Q9 but to set the quiescent current
through the output transistors. We’ll
discuss how this is done in the setting
up procedure.
It is important that the bias voltage
produced by Q9 tracks the temperature of the output stage transistors. As
the output transistors become hotter,
Q9’s collector-emitter voltage should
drop, so that the quiescent current is
reduced and the danger of thermal
runaway is averted. Our prototype
photo this month shows Q9 directly
on top of Q12 but next month it will
be shown above Q12.
Output stage
The output stage of the amplifier is
effectively a complementary symme-
try emitter follower, comprising six
NPN transistors and six PNP transistors. We need this many transistors to
safely deliver the high peak currents
involved (up to 17 amps peak) at high
voltages. The load line curves of Fig.2
demonstrate that while 12 output
transistors are adequate to cope with
reactive 4Ω loads (typified by the
2.83Ω + j2.83Ω curve), there is not a lot
of power capacity to spare when you
look at the 1200W and SOAR hyperbola curves. In other words, while 12
big power transistors might look like
a lot, every one of them is needed to
safely deliver full power into typical
4Ω loudspeaker loads.
Each output power transistor has a
0.47Ω emitter resistor and this more
or less forces the output transistors to
roughly share the load currents. If one
of the power transistors tends to take
more than its share of load current, the
corresponding voltage drop across its
emitter resistor will be proportionately higher and this tends to throttle the
transistor back until its current comes
back into line with the others.
The emitter resistors also help to
stabilise the quiescent current to a
small degree and they slightly im-
prove the frequency response of the
output stage by providing current
feedback.
Gain setting
Negative feedback is applied from
the output stage back to the base of
Q2 via an 18kΩ resistor. The amount
of feedback is set by the 18kΩ resistor
and the 560Ω resistor at the base Q2.
These set the gain of the amplifier to
33. The low frequency rolloff is set
mainly by the ratio of the 560Ω resistor
to the impedance of the 100µF capacitor. This gives a -3dB point of about
2.8Hz. The 2.2µF input capacitor and
18kΩ bias resistor to Q1 have similar
effect and give a -3dB point of 4Hz.
The two time-constants combined
give an overall rolloff of about 7Hz.
At the high frequency end, the
820pF capacitor and 1.2kΩ resistor
feeding the base of Q2 form a low pass
filter which rolls off frequencies above
160kHz (-3dB). The overall amplifier
frequency response is demonstrated
in the curves of Fig.7 and Fig.8.
An output RLC filter comprising
a 5.7µH choke, a 6Ω resistor and a
0.15µF capacitor couples the signal
to the loudspeaker. It isolates the am-
SILICON
CHIP
This advertisment is now out of date.
Please feel free to visit the advertiser’s website:
www.emona.com.au
August 1997 31
Parts List For 500W Amplifier Module
500 amplifier PC board
1 PC board, code 01208971,
362mm x 99mm
4 20mm fuse clips
2 5A or 7.5A 20mm fuses (see
text)
1 coil former, 24mm OD x
13.7mm ID x 12.8mm long,
(Philips 4322 021 30362)
1 2-metre length 1mm
enamelled copper wire
1 200Ω trimpot (Bourns 3296W
or similar) (VR2)
1 100Ω multi-turn horizontal
mount trimpot (VR1)
7 PC stakes
2 TO126 heatsinks, Jaycar Cat.
HH8504 or similar
1 single-sided heatsink, 400mm
wide x 118mm high x 48mm
deep, or two 200mm x 118mm
x 48mm
14 TO-3P insulating washers
2 TO-126 insulating washers
17 3mm x 10mm screws
3 3mm nuts
Semiconductors
2 2N5401 PNP transistors (Q1,Q2)
2 BC556 PNP transistors (Q3,Q25)
4 BC546 NPN transistors
(Q4,Q5,Q7,Q24)
1 MJE350 PNP transistor (Q6)
2 MJE340 NPN transistors
(Q8,Q9)
7 MJL21194 NPN power
transistors (Q10,Q12-Q17)
plifier from any large capacitive react
ances in the load and thus ensures
stability. Perhaps more importantly,
the filter attenuates any RF signals
picked up by the speaker leads and
stops them being fed back to the amplifier’s input stage where they could
cause audible breakthrough – no-one
likes listening to radio stations when
they are supposed to be hearing CDs.
Overload protection
& offset adjustment
Two other circuit features need to be
mentioned: DC offset adjustment and
overload protection. Strictly speaking,
the DC offset adjustment is not really
necessary if the amplifier is not to be
used with an output transformer, as
32 Silicon Chip
7 MJL21193 PNP power
transistors (Q11,Q18-Q23)
4 1N914 small signal diodes
(D1,D2,D3,D4)
2 1N4936 fast recovery diodes
(D5, D6)
1 BZX55C3V3 3.3V 0.5W zener
diode (ZD1)
Capacitors
4 100µF 100VW electrolytic
1 100µF 16VW electrolytic
1 2.2µF 16VW electrolytic
1 0.15µF 275VAC (Philips MKP
2222 336 10154)
5 0.1µF 100VW MKT polyester
1 820pF MKT polyester or
ceramic
1 100pF 500V ceramic (Philips
2222 655 03101)
Resistors (0.25W, 1%)
4 22kΩ 1W
2 18kΩ
1 6.8kΩ 1W
1 1.2kΩ
1 560Ω
1 470Ω
2 390Ω 5W 10%
4 270Ω
2 220Ω
1 180Ω
1 120Ω
5 100Ω
2 30Ω
3 18Ω 1W
12 0.47Ω 5W 10%
it would be if it was driving a 100V
line transformer for PA work. However, because we envisage that some
readers will want to use the amplifier
for public address, we have included
DC offset adjustment.
This is provided by the 100Ω trim
pot (VR1) between the emitters of the
input pair, Q1 & Q2. VR1 is used to
adjust the current balance between the
input pair and this, because it is a DC
feedback circuit, causes the DC offset
at the output to vary. The trimpot is
adjusted to make the DC offset as close
to 0V as possible; it should be possible
to keep to less than ±5mV.
Transistors Q24 & Q25 and diodes
D3 & D4 provide the overload protection feature. Q24 monitors the current
flow through the emitter resistor of
Q12, via a voltage divider consisting
of a 300Ω resistor and a 270Ω resistor.
Normally, Q24 & Q25 are off and
play no part in the amplifier’s operation. However, if the current through
the 0.47Ω resistor of Q12 exceeds
about 3 amps, Q24 begins to turn on
and shunts the base current from Q10,
the associated driver transistor. This
means that not only is Q12 throttled
back, but so are the other five NPN
output transistors, Q13-Q17, because
they all must operate in an identical
way. Hence the peak output current
is prevented from exceeding about
18 amps. This means the amplifier
can deliver full power into a 4Ω load
but if a 2Ω load, for example, was
connected, the power output would
be heavily limited.
The same process happens for Q25
which monitors the emitter current of
Q18 (and thus Q19-Q23). The diodes
D3 & D4 are included to prevent Q24
& Q25 from shunting the drive signal
when they are reverse-biased; this
happens for every half cycle of the
signal to the driver transistors.
Diodes D5 & D6 are included as
part of the protection circuitry and
they absorb any large spikes which
may be generated by the inductance
of the loudspeaker when the current
limiting circuit cuts the drive to the
output transistors. D5 & D6 are fast
recovery diodes, included to ensure
their operation at high frequencies
and high power.
Thermal cutout
Because the overload protection
simply limits the current to the load,
the output transistors and the fuses
are protected from sudden death in
the case of a momentary short circuit
but if the overload (or short circuit) is
maintained and the drive continues,
the amplifier will very rapidly overheat and may still expire unless the
fault condition is correctly quickly.
To prevent failure of the output transistors, the circuit includes an 80°C
thermal cutout. This is not shown on
the circuit of Fig.1 but is a vital part
nevertheless. It is part of the relay
protection circuit to be presented
next month.
Next month, we’ll present the circuit of the power supply and for the
relay protection circuit and give the
construction details of the complete
SC
amplifier.
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