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Studio 350 Pow
Amplifier Modu
Want an audio power amplifier with some real grunt?
Want an audio power amplifier which is really quiet
and has very low distortion? Here is the one answer for
both desires. The Studio 350 is a rugged power amplifier
module capable of delivering 200 RMS watts into an
8-ohm load and 350 watts RMS into a 4-ohm load, at
very low distortion.
Pt.1: By LEO SIMPSON & PETER SMITH
F
OLLOWING THE outstanding
success of our SC480 power amplifier module published in the
January & February 2003 issues, we’ve
taken the lessons learned there and
from our Ultra-LD series published in
2000 and 2001 and applied them to a
much bigger power amplifier.
There is no doubt the publication of
the SC480 triggered off a lot of interest
and since then we’ve had readers suggesting we update the 300W amplifier
from the February 1980 issue of ETI.
Others have asked about the possibility of upgrading the SC480 with bigger
transistors and higher supply rails or
variations on that theme. So the seeds
were sown. A bigger amplifier was
called for. But how much bigger? And
using which transistors?
Looking at the SC480, for example,
you can’t increase the power output
by simply substituting bigger output
transistors and increasing the supply
rails to some likely value. If you were
12 Silicon Chip
to take that approach, other transistors
in the circuit would blow up. And
if you’re driving low impedance (ie,
4-ohm) loads, the output transistors
could easily expire as well.
Our first approach was to decide
on the target power output, given a
likely supply rail. Given that we have
already published amplifiers capable
of delivering 100 watts into 8-ohm
loads (ie, the Ultra-LD series), the next
likely step would be to aim for 200
watts into an 8-ohm load. A few backof-an-envelope calculations show that
we would need supply rails of about
±70V or a total of 140V.
Naturally, we would also want to
drive 4-ohms loads and with those
same supply rails we would expect
to obtain around 350 watts. But how
many output transistors and what type
would be required? As you can see
from the photos and circuit, we have
used eight 250V 200W plastic power
transistors: four MJL21193/4 comple-
mentary pairs. These are teamed with
the high-performance MJL15030/31
complementary driver transistors.
In addition, we have used some new
high-voltage low-noise transistors in
the input stage and highly linear highvoltage video transistors in the voltage
amplifier stage. In other respects, the
amplifier circuit is not much different from that of the SC480. Equally
important, we have used the same PC
board distortion-cancelling topology
as in the SC480. The net result is a
rugged power amplifier with very low
residual noise and distortion.
Load lines and power ratings
So why did we end up using eight
200W transistors in order to get just
200W into 8Ω and 350W into 4Ω?
It might seem like over-kill but it is
not. To work out the dissipation in a
transistor, you need to draw the load
lines. These show power dissipation
in the active device (in this case, one
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er
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half of the output stage, consisting of
four transistors). The vertical axis is
in Amps while the horizontal axis is
Volts. The various load lines for our
amplifier are shown in Fig.1.
For a start, we plotted the lines for
8-ohm and 4-ohm resistive loads and
these are straight lines, showing all
possible conditions. The two resistive lines start at the 70V mark on the
horizontal axis, corresponding to the
supply voltage applied across one half
of the output stage (either the NPN or
the PNP transistors). For the 4Ω load,
the load line runs up to 17.5A on the
vertical axis, corresponding to the current delivered if the active device was
fully turned on (ie, 70V ÷ 4).
Similarly, for an 8Ω load, the load
line runs up to 8.75A on the vertical
axis (ie, 70V ÷ 8). These load lines
show the instantaneous power dissipation at any possible signal condition
(including an output short circuit).
Also shown on the diagram are two
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hyperbolas. One represents the maximum safe power
(for one second!) dissipation of four
parallel-connected MJL21193/94 tran
-sistors. Depending on the instantaneous voltage across the transistors,
this can be more than 900W for low
voltages, reducing to 720W at 80V, and
ultimately to just 400W at 250V (not
shown on the curve). This hyperbola
represents the maximum dissipation
the four transistors can withstand under a non-repetitive one-second pulse,
the so-called “Safe Operating Area”.
Since the resistive load lines are
well below the one-second SOA hyperbola, you may think that the transistors are operating far below their
maximum ratings and so they would
be, if all they had to drive was resistive loads. Sadly, loudspeakers are not
resistive; they can be resistive,
inductive or capacitive, depending on the signal frequency. Usually
they are inductive which means the
load current lags the load voltage.
This has two effects. First, the voltage across the output transistors can go
much higher than the half-supply value
of 70V. Conceivably, it can run to the
full supply voltage of 140V (or beyond,
if driven into clipping on an inductive
load). Second, the instantaneous power
dissipation across the power transistors
can go far in excess of that shown for a
resistive load line.
To show this effect, we have drawn
8Ω and 4Ω reactive load lines which
represent speakers with complex impedances of 5.6Ω + j5.6Ω and 2.83Ω
+ j2.83Ω, respectively. In the 8Ω
case, the 5.6Ω represents the voice
coil resistance while the j.5.6Ω is the
coil inductance. The resulting curved
load lines extend well beyond 70V (to
almost 110V) and also show instantaJanuary 2004 13
Fig.1: this diagram
shows the resistive
and reactive load
lines for both 4Ω and
8Ω loads. Also shown
are two hyperbolas.
The blue curve shows
the maximum safe
operating area of four
parallel-connected
MJL21193/MJL21194
transistors, while
the red curve shows
the derated power
curve for 50°C case
temperature.
neous dissipation figures far in excess
of that for the resistive load lines. In
fact, you can see that in the case of the
4Ω reactive case, there is far less power
margin to spare.
In fact, we have also drawn the derated power hyperbola (50°C) for four
transistors on Fig.1 and as you can
see, it touches the 4Ω reactive curve.
Does this mean there is a problem?
Well no, because the load lines show
instantaneous power dissipation, not
average or total power dissipation. As
long as the load lines are below the
SOA curve, everything is OK.
All of the foregoing is a shortened
explanation of the process whereby
we decided to use eight transistors. It
shows that eight is a good conservative
figure whereas six of these transistors
would not be enough.
Finally, before we leave the discussion on load lines, we need to mention
short circuit and overload protection.
Apart from fuses, this amplifier circuit
has no protection. We could have chosen to run with six power transistors
if we had incorporated “load line”
protection into the circuit. This uses
a pair of transistors to monitor the
output transistor voltage and current
conditions and then limit the base
drive signal when the load line is
exceeded.
Such circuits can work quite well to
protect the output stage but in practice
their rapid switching action causes a
burst of high frequency oscillation to
be superimposed on the output signal.
This means that not only do you get
Fig.2: total harmonic distortion versus power at 1kHz into
an 8-ohm load (10Hz-22kHz measurement bandwidth).
14 Silicon Chip
horrible distortion but the amplitude
of the burst can be enough to overload
and burn out tweeters if the overdrive
situation persists.
Therefore, while we regard load
line protection as important for PA
amplifiers (which can easily have their
output leads shorted), it is not desirable for a hifi amplifier. If you do short
the outputs of this amplifier when it
is under full drive, there will be a big
spark and hopefully the only thing
to be damaged will be the 5A fuses.
If the fuses were increased in rating,
the amplifier could ostensibly drive a
2Ω resistive load without damage, so
we think the 5A fuses should provide
adequate short circuit protection. Oh,
but we don’t recommend driving a
2Ω load!
Fig.3: total harmonic distortion versus power at 1kHz into
a 4-ohm load (10Hz-22kHz measurement bandwidth).
www.siliconchip.com.au
Fig.4: harmonic distortion versus frequency at 160W into
an 8-ohm load (22Hz-80kHz measurement bandwidth).
By the way, we strongly recommend
the use of a relay protection circuit
to prevent loudspeaker damage in
the event of a catastrophic fault in
the amplifier. A suitable circuit was
featured in the October 1997 issue of
SILICON CHIP.
Amplifier module
Two versions of this amplifier module are possible, both using the same
PC board pattern. The one presented
here employs a cast aluminium heatsink with an integral shelf which is
convenient for mounting the power
transistors. This heatsink is 300mm
wide and the PC board itself is 240
x 136mm so the overall assembly is
quite large.
The alternative approach is to
mount the output transistors vertically
on a single-sided or fan heatsink, in
which case the PC board would be
trimmed to 240mm wide by 100mm
deep. This latter approach takes up
less chassis space. Both approaches
will be described in the constructional
details to be presented next month.
Performance
As already noted, the Studio 350
delivers up to 200W RMS into an
8-ohm load and up to 350W into a
4-ohm load. Music power figures are
substantially higher, around 240W
into an 8-ohm load and 480W into a
4-ohm load. These figures apply only
for the suggested power supply which
we will come to later.
Fig.2 shows the total harmonic
distortion versus power at 1kHz into
an 8-ohm load while Fig.3 shows
distortion versus power at 1kHz into
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Fig.5: distortion versus frequency at 250W into a 4-ohm
load (22Hz-80kHz measurement bandwidth).
a 4-ohm load. As you can see, for
an 8-ohm load, distortion is around
.002% or less up to about 180W, rising to around .03% or thereabouts at
200W. At low powers, below 0.5W,
the distortion figure rises but that is
due to residual noise, not distortion.
In reality, at low powers the distortion
is well below .001%.
Similarly, for a 4-ohm load, distortion is around .0045% or less for powers up to around 280W, rising to 0.1%
at around 350W. These figures were
taken with a measurement bandwidth
of 22Hz to 22kHz.
Fig.4 shows harmonic distortion
versus frequency at 160W into an
8-ohm load while Fig.5 shows distortion versus frequency at 250W into a
4-ohm load. Both these curves were
taken with a measurement bandwidth
of 22Hz to 80kHz.
All of these distortion curves show
a performance which is outstanding.
For 8-ohm loads, it is very close to that
of the Ultra-LD amplifier published
in November 2001, December 2001
and January 2002. As well, it is better than our Plastic Power amplifier
of April 1996 and far better than our
500W amplifier described in August,
September and October 1997. That’s
progress!
This amplifier is also extremely
quiet: -122dB unweighted (22Hz to
22kHz) or -125dB A-weighted. This is
far quieter than any CD player!
Fig.6 shows the frequency response
at 1W into 8Ω. It is 1dB down at 15Hz
and 60kHz.
Circuit description
The full circuit is shown in Fig.7
and employs 15 transistors and five
diodes. In essence, it is quite similar
in layout to the SC480 design referred
to earlier, which was based on a design
originally produced by Hitachi.
The input signal is coupled via a 1µF
bipolar capacitor and 2.2kΩ resistor
Fig.6: this graph
shows the frequency response at
1W into 8Ω. It is
just 1dB down at
15Hz and 60kHz
and is virtually
flat between those
frequencies.
January 2004 15
16 Silicon Chip
www.siliconchip.com.au
Performance
Fig.7: the circuit uses eight audio
output transistors to give a rugged
design with low distortion. The
voltage readings on the circuit were
taken with no input signal.
to the base of Q2. Q2 & Q3 are a differential pair using Hitachi 2SA1084
low-noise transistors which have a
collector-emitter voltage rating of 90V,
necessary because we are using 70V
rails. Transistor Q1 and diodes D1 &
D2 make up a constant current source
running at about 1mA to set the current through the differential pair at
0.5mA each.
Trimpot VR1 in the emitter circuit
to the differential pair is provided to
adjust the offset voltage and thereby
trim the output DC voltage very close
to 0V (within a millivolt or so). This
is largely academic if you are driving
normal 4-ohm or 8-ohm loudspeakers but is particularly desirable if you
intend driving electrostatic speakers
which usually have a high voltage
step-up transformer with very low
primary resistance.
The same comment applies if the
amplifier is used to drive 100V line
transformers. Just to explain that, if you
have a transformer primary resistance
of 0.1Ω and a DC output offset from the
amplifier of just 20mV, the resulting
current through the transformer will be
200mA! Not only will this magnetise
the core and degrade the transformer’s
performance, it will also result in additional power dissipation of 14W in
one half of the amplifier’s output stage.
This is not good! Hence, trimpot VR1
has been included.
Signals from Q2 & Q3 drive another
differential pair, Q4 & Q5, which have
Output Power . . . . . . . . . . . 200W into 8Ω; 350W into 4Ω
Music Power . . . . . . . . . . . 240W into 8Ω; 480W into 4Ω
Frequency Response . . . . . -1dB at 15Hz and 60kHz at 1W (see Fig.6)
Input Sensitivity . . . . . . . . . 1.75V for 200W into 8Ω
Harmonic Distortion . . . . . . Typically .002% at normal listening levels
. . . . . . . . . . . . . . . . . . . . . . (see graphs)
Signal-to-Noise Ratio . . . . . -122dB unweighted (22Hz to 22kHz); -125dB
. . . . . . . . . . . . . . . . . . . . . . A-weighted, both with respect to 200W into 8Ω
Damping Factor . . . . . . . . . 75 at 10kHz, with respect to 8Ω
Protection . . . . . . . . . . . . . . 5A supply fuses (see text)
Stability. . . . . . . . . . . . . . . . Unconditional
a “current mirror” as their collector
loads. The current mirror comprises
diode D3 and Q6, essentially a variation of a constant current load which
ensures high linearity in Q5. Q4, Q5
and Q6 are BF469 and BF470 types
which are high-voltage (250V) video
transistors, selected for their excellent
linearity and wide bandwidth (Ft is
60MHz).
Q7 is a “Vbe multiplier”, so-called
because it multiplies the voltage between its base emitter to provide a
floating voltage reference to bias the
output stage and set the quiescent
current. Quiescent current is needed
in all class-B amplifiers, to minimise
crossover distortion. In fact, this amplifier displays no trace of crossover
distortion.
We use an MJE340 transistor for Q7
even though a small signal transistor could easily handle the task. The
reason for using a power transistor is
that its package and junction does a
better job of tracking the temperature
dependent changes in the junctions
of the output power transistors and
thereby gives better overall quiescent
current control.
The driver transistors are the high
performance MJE15030 and MJE15031
made by On Semiconductor (previously Motorola). These were first used
DANGER: HIGH VOLTAGE!
The 100VAC from the transformer secondaries and the 140V DC supply
across the filter capacitor bank and the amplifier supply rails is potentially lethal! After the power supply wiring is complete and before you
apply power, mount a clear Perspex sheet over the capacitor bank to
protect against inadvertent contact – now or in the future! Note that the
capacitors take some time to discharge after the power is switched off.
Fig.8: the power supply uses a 50V-0-50V transformer
to drive a 35A bridge rectifier and two banks of three
8000µF 75V capacitors to develop supply rails of ±70V.
www.siliconchip.com.au
January 2004 17
This view shows the fully completed audio amplifier module. The construction details are in next month’s issue.
by us in the Ultra-LD series and have
a minimum current gain-bandwidth
product (Ft) of 30MHz. These drive the
paralleled output stage MJL21193/94
transistors which themselves have a
typical Ft of around 6MHz.
Overall, this is a far superior line-up
of transistors to that used in the SC480
amplifier (January & February 2003)
and it results in far better distortion
performance at high power and at
higher frequencies.
Each of the power transistors in the
output stage has 5W wirewound emitter resistor of 0.47Ω. This relatively
high value has the disadvantage that it
causes a slight reduction in power output but this has been done to provide
improved current sharing between
the output transistors – an important
factor in a high-power design.
Although not shown in the photographs, one of our prototypes used
non-inductive wirewound emitter
resistors. These have been recommended in some past designs in overseas magazines, in order to minimise
secondary crossover distortion effects.
Our tests showed no benefit in this design (probably because of the PC board
layout) and so they are not specified –
ordinary wirewound emitter resistors
are OK in this design.
Two 1N4936 fast recovery diodes
18 Silicon Chip
are reverse-connected across the
output stage transistors. Normally,
these do nothing but if the amplifier
is driven into clipping when driving
highly inductive speakers or transformers, the diodes safely clamp the
resulting back-EMF spikes to the
supply rails.
Negative feedback
Overall negative feedback is applied
from the output stage via the 22kΩ
resistor to the base of Q3. The voltage
gain is set by the ratio of the 22kΩ
resistor to the 1kΩ resistor also connected to the base of Q3. This gives a
voltage gain of 23 (+27dB). The 47µF
bipolar capacitor in series with the
1kΩ resistor sets the -3dB point of
the frequency response to about 3Hz.
The other factor in the amplifier’s low
frequency response is the 1µF bipolar
input capacitor.
We have used non-polarised capacitors for the input and feedback
coupling instead of conventional
electrolytic capacitors because the
low voltages present in this part of
the circuit are insufficient to polarise
conventional electros. Incidentally,
some readers may disagree with our
choice of electros in the signal path but
the alternative of plastic dielectric capacitors is not very attractive; they are
large and expensive and unavailable,
in the case of 47µF. Nor do we think
that electrolytic capacitors, properly
used, are the cause of high distortion
in audio circuits; there’s no evidence
of it in the case of this circuit.
The 330pF shunt capacitor and
2.2kΩ resistor in series with the input
signal constitute an RC low-pass filter,
rolling off the high frequencies above
200kHz. The 68pF capacitor between
Q5’s base and emitter rolls off the
open loop gain to ensure stability with
feedback applied.
Note that this capacitor can be ceramic or polystyrene but must have
a rating of 250V. This is because the
signal at this part of the circuit can be
as high as 45V RMS (127V peak-topeak). Other capacitor types (such as
monolithics) are definitely not recommended.
As in our previous amplifiers, the
output signal to the loudspeaker is
fed via an RLC filter consisting of
6.8µH choke, a 6.8Ω wirewound
resistor and a 150nF capacitor. This
very well-proven filter network was
originally developed by Neville Thiele
and published in the September 1975
issue of the “Proceedings of the IREE”.
The filter has two benefits: ensuring
stability of the amplifier with reactive
loads and as an attenuator of RF and
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Scope1: this waveform shows the excellent square wave
response of the amplifier, taken at 1kHz and 102V p-p into
8Ω. This equates to a power output of about 300W RMS.
Scope2: these waveforms show a 150W sinewave at 1kHz
and the resulting total harmonic distortion waveform (ie,
noise and distortion) which is at about .0015%.
Scope3: this is the pulse waveform used to measure music
power. Note the excellent stability of the amplifier,
particularly the recovery after the pulse.
Scope4: the same waveform as in Scope3 but with the
scope switched to a faster timebase. In this case, the
amplifier is delivering over 240W RMS into an 8-ohm load.
mains-interference signals which are
inevitably picked up by long loudspeaker leads.
Power supply
Fig.8 shows the power supply and
as you can see, we’ve “gone for the
doctor” on this one. It’s a vital part of
the performance package and unfortunately, with all those big electrolytic
capacitors, is likely to be more expensive than the module itself. The consolation is that the same power supply
could be used for a stereo version with
two amplifier modules, provided the
power transformer was uprated.
The 500VA transformer used has
two 50V windings which are connected together to form a centre tap. This
transformer drives a 35A bridge rectifier and two banks of three 8000µF 75V
www.siliconchip.com.au
capacitors to develop ±70V supply
rails. The 470nF capacitors are used
to provide high frequency bypassing,
while the 15kΩ 1W resistors are used
as “bleeders” across the electrolytic
capacitors.
PC board topology
Finally, as noted at the start, the PC
board has been laid out using the same
distortion-cancelling topology used in
the SC480. It also has “star” earthing
whereby all earth currents come back
to a single point on the board. This
careful separation of output, supply
and bypass currents avoids any interference with the signal currents and
the distortion that this could cause.
As far as the “distortion cancelling”
technique is concerned, this involves
laying the copper tracks so that the
magnetic fields produced by the asymmetric currents in the output stage are
cancelled out, as far as possible. These
asymmetric currents (think of them as
half-wave rectified output signals) and
their resultant magnetic fields induce
unwanted distortion signals into the
input stage involving Q2 & Q3.
Arguably, the field cancelling technique is not quite as successful in this
design as in the SC480, because this
new PC board is much larger and the
output devices are more spread out.
Even so, it is very worthwhile and constructors will not have to worry about
whether the performance of their module is as good as the prototype featured
here. As long as you closely follow
the wiring layout in the construction
article next month, you can expect the
SC
results to be very good.
January 2004 19
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