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If you can’t afford a high-performance amplifier and loudspeakers, you can still have the best possible hifi sound, with this headphone amplifier and a set of high-quality headphones. By NICHOLAS VINEN Hifi Stereo Headphone Amplifier, Pt.1 Y ES, WE KNOW that the UltraLD amplifier modules described elsewhere in this issue are “over the top” for many people, especially those living in small home units and those who have to worry about sound levels annoying their neighbours. But why not listen via a good pair 62  Silicon Chip of headphones? Spend a few minutes looking around the internet and you will find all manner of hifi headphone amplifiers that claim to have top-notch performance. In most cases, there is little or no performance data to prove it. Before spending upwards of $1000 on a headphone amplifier we’d want to know just how good it is! Our new headphone amplifier has a performance virtually the same as our benchmark 20W Class A Stereo Amplifier (May-September 2007). Its distortion at 100mW is lower than that from even the best CD and BluRay players. So essentially what you hear siliconchip.com.au CON1 LEFT INPUT INPUT RF FILTERING PREAMPLIFIER (GAIN = 0 TO –15) POWER AMPLIFIER (GAIN = –1) OUTPUT RLC FILTER CON4 HEADPHONE OUTPUT CON2 RIGHT INPUT Fig.1: this block diagram shows the basic arrangement of the headphone amplifier. It incorporates RF filtering, a stereo preamplifier, stereo amplifier, output isolation filters and a regulated power supply. is what is recorded on the CD – no more and no less. This project does not supersede the Portable Headphone Amplifier for MP3 Players (April 2011) since that one is small, light and batterypowered. That design was intended for use “on the go” and to give much better sound than normally available from iPods and MP3 players. This new headphone amplifier will also drive 8Ω loudspeakers and has a music power of 4.25W for both channels driven. This is more than adequate if you have reasonably efficient loudspeakers in your study, office or bedroom. It is housed in a half-size 1U steel case just 210mm wide, 49mm high and 125mm deep and is powered by an AC plugpack (no 230VAC mains wiring). The interior of the case is filled by the PCB which accommodates all the components. There is no other wiring to do; just assemble the PCB, fit it into the case and you’re finished. Circuit features Fig.1 shows the block diagram of the unit, while Fig.2 shows the complete circuit. It looks huge, doesn’t it? That’s partly because it shows both channels. It can be split into two sections, with the preamplifiers and power supply on the lefthand side and the power amplifiers on the righthand side. The preamplifier for each channel is based on three op amps so three LM833 dual op amps are used. The preamp configuration is a classic Baxandall siliconchip.com.au design. The preamplifier is inverting and has a gain range from zero to -15. The reason for such a wide range in gain is that we have to provide for a large variety of headphone impedances and sensitivities. 8Ω headphones require a much lower voltage swing for the same power compared to 600Ω phones. Driving 8Ω headphones from a CD player (typically 2V RMS) may require a gain of 0.25 or less while using 600Ω phones with a line level signal (0.775V RMS or sometimes less) could require a gain of several times. The Baxandall preamplifier circuit has the advantage that it varies its gain according to the setting of potentiome- ter VR1. As a result, the residual noise level is kept low at the low gain settings most commonly required. Like a traditional preamplifier, its gain can go all the way down to zero and up to some fixed number, in this case, 15. Another advantage of this circuit is its log-like gain curve from a linear potentiometer, which generally have superior tracking compared to log pots. All but the most expensive “log” law potentiometers actually use a dual linear taper and so they don’t really have an accurate log response either. The two power amplifiers on the righthand side of the circuit are loosely based on the 20W Class-A Amplifier Features & Specifications Main Features • • • • • Suits 8Ω – 600Ω headphones and ear-buds Very low distortion and noise Plugpack-powered (no mains wiring) Short-circuit protected Can also drive efficient 8Ω loudspeakers Specifications (Figs.3-7) Rated power: 100mW (8-100Ω), 25mW (600Ω) THD: 0.0006% <at> 1kHz; 20Hz-22kHz bandwidth Signal-to-noise ratio: -113dB unweighted; 20Hz-22kHz Frequency response: ±0.15dB, 20Hz-20kHz Channel separation: -73dB <at> 1kHz Maximum power: 4.25W (8Ω), 3W (16Ω), 1.5W (32Ω), 800mW (60Ω), 80mW (600Ω) Class-A power: 18mW (8Ω), 36mW (16Ω), 72mW (32Ω), 80mW (600Ω) Music power: 4.25W into 8Ω, both channels driven (see text) September 2011  63 10 +12V K D9 1N4004 100nF K D15 BAT42 A LEFT INPUT A CON1 L1 470nF 100 8 3 2 100pF NP0 100k +11.8V –11.8V 220 F 1 IC1a VR1b 10k LIN 100pF NP0 100k 22 F K 10k 3 D16 BAT42 A 2 8 IC2a 680 1 6 22k 7 IC2b 5 –11.8V 220 F 4 IC1, IC2, IC3: LM833 +11.8V K VOLUME RIGHT INPUT D17 BAT42 CON2 L2 A 470nF 100 5 6 100pF NP0 100k 220 F 100nF 220 F 7 IC1b 100nF –11.8V VR1a 10k LIN 100pF NP0 4 100k 22 F K D18 BAT42 A 10k 3 –11.8V 2 8 IC3a 680 1 6 5 22k 1k 7 IC3b 220 F 4 D10 1N4004 K –11.8V A D3 1N4004 10nF F1 1A FAST* K D1 1N4004 REG1 7812 K A IN 12V AC INPUT 2200 F* +12V OUT K GND 10nF 10 A 100nF POWER D4 1N4004 220 F A CON3 S1 A K GND IN A OUT A  LED1 D6 1N4004 100nF 22k K K 2200 F* D2 1N4004 +12V 220 F –12V 30k –12V REG2 7912 10nF A K D5 1N4004 SC 2011 * FOR DRIVING SPEAKERS, INCREASE THE RATING OF F1 TO 2A (FAST) AND ALSO INCREASE THE VALUE OF THE TWO 2200 F CAPACITORS TO 4700 F (SEE TEXT) HI-FI STEREO HEADPHONE AMPLIFIER Fig.2: the complete circuit of the Hifi Stereo Headphone Amplifier. The stereo preamplifier section is at upper left and is based on three low-noise dual op amps (IC1-IC3). This stage provides a variable gain of 0-15 depending on the setting of VR1 which functions as the volume control. The two identical power amplifiers are shown at right and these drive the headphones via RLC filters (for stability) and a 6.35mm jack socket. The linear regulated power supply is at lower left and this derives regulated ±12V rails from a 12V AC plugpack. 64  Silicon Chip siliconchip.com.au 10 K D11 1N4004 220 A Q5 BC559 E 47 F Q6 BC559 2.2k B E C C 43 2.2k B +12V E B 220 F Q7 BC559 –12V C 10k 22 C B 1.1k 220 F 10k 100 100nF 910 VR2 500 100 E E C C Q2 Q1 BC559 BC559 B E 2.2k 47 F TP1 + C Q10 B BD139 1.2 E 1.2 TP2 – 1.8k B Q11 TIP31 + 680pF NP0 A 28.5mV 1.2 220pF NP0 10k 22 E B E Q3 BC549 C E D12 1N4004 K C B B E B Q8 BC549 E 68 Q4 BC549 C Q12 TIP32 2.2k C B E 150nF HEADPHONE SOCKET 47 –12V 10 220 A Q17 BC559 E 47 F Q18 BC559 2.2k B E E B 220 F 150nF Q19 BC559 –12V C 10k 22 C B 1.1k 220 F 100 47 F 910 100nF E C C Q14 Q13 BC559 BC559 B VR3 500 100 E E 2.2k 10k Q23 TIP31 TP3 + C Q22 B BD139 1.2 E 1.2 1.8k B K B 1.8k 10k Q20 BC549 B 22 B E Q15 BC549 C D14 1N4004 K B E B C E 68 Q16 BC549 2.2k E E C Q24 TIP32 Q26 BC328 7812 C B C E GND IN OUT GND Q21 BC338 7912 –12V 2.2k 68 D8 1N4004 1.2 – C 10 TP4 – A 28.5mV 1.2 220pF NP0 L4 4.7 H 28.5mV + 680pF NP0 CON4 +12V 43 2.2k B C C 10 Q9 BC338 2.2k 68 D13 1N4004 Q25 BC328 C A K L3 4.7 H K 1.8k C D7 1N4004 1.2 – B 28.5mV GND 47 A IN OUT IN BD139 D1–D14: 1N4004 A siliconchip.com.au K LED1 D15–D18: BAT42 A K K A TIP31, TIP32 BC328, BC338, BC549, BC559 B E B C C B E C C E September 2011  65 0.01 THD+N vs Frequency, 100mW, 20Hz-80kHz Bandwidth 0.01 THD+N vs Frequency, 100mW, 20Hz-22kHz Bandwidth 0.002 0.001 0.0005 0.0002 0.002 0.001 0.0005 0.0002 0.0001 20 50 100 200 500 1k 2k 5k 10k 0.0001 20 20k 50 100 200 Frequency (Hz) Frequency Response, 100mW -50 -60 +1.5 -65 +1 -70 +0.5 -75 -0 -0.5 -1.5 -95 -2 -100 -2.5 -105 100 200 500 1k 2k 5k 10k 20k 50k 100k 06/10/11 14:37:19 -110 20 50 100 200 500 1k 2k 5k 10k 20k Frequency (Hz) Fig.5: the frequency response for typical loads. The lowend -3dB point is around 3Hz, while the high-frequency response is defined by the output filter and so varies with load impedance. This results in a slight treble boost for loads of 16Ω and above. 66  Silicon Chip 20k 8 16 32 600 Frequency (Hz) but with smaller output transistors and heatsinks. The power amplifiers invert the signal again, so the unit’s outputs and inputs are in-phase. Since there is so much gain available in the preamps, the power amplifiers operate at unity gain (ie, -1). This improves the noise performance and maximises the feedback factor, keeping distortion exceedingly low even with run-of-themill output transistors. Because the headphone connector is a jack socket, the outputs can be briefly short circuited if the plug is inserted or removed during operation. As a result, the design incorporates short-circuit 10k -85 -90 50 5k -80 -1 20 2k Channel Separation vs Frequency, 100mW -55 Crosstalk (dBr) Amplitude Variation (dBr) 06/10/11 14:14:52 +2 -3 10 1k Fig.4: THD+N but with a 22kHz upper bandwidth limit. This gives more accurate figures for low frequencies but also eliminates high-frequency signal harmonics, hence the artificial drop in distortion above 7kHz. 8 16/32 600/100k +2.5 500 Frequency (Hz) Fig.3: total harmonic noise and distortion (THD+N) vs frequency for four typical load impedances. The slight increase in distortion above 3kHz for a 600Ω load is due to slew rate limiting. +3 06/10/11 13:27:03 8 16 32 600 (25mW) 0.005 Total Harmonic Distortion + Noise (%) 0.005 Total Harmonic Distortion + Noise (%) 06/10/11 13:27:03 8 16 32 600 (25mW) Fig.6: channel separation versus frequency. Most of the crosstalk that occurs is due to shared ground paths; it is resistive and so constant with frequency but varies with load impedance. Above 5kHz, some additional capacitive and inductive crosstalk is apparent. protection to prevent any damage. Our noise and distortion figures are quoted at 100mW for 8-32Ω and 25mW for 600Ω. With efficient headphones, this is enough to generate very high sound levels. For most headphones, a typical listening level is 0.5-5mW. Common mode distortion By lowering the gain, we get a higher feedback factor (which is good) but we also increase the possibility of common-mode distortion. This can reduce the effectiveness of a high feedback factor so that the distortion reduction (due to the feedback) is not as much as would otherwise be the case. While the differential input voltage (ie, the voltage between the two inputs) of an amplifier operating in closed loop mode is very small, both input voltages can still have large swings, especially when the amplifier is being driven hard. This is the “common mode” signal, ie, signal common to both inputs. For a non-inverting amplifier, the common mode voltage is the output voltage swing divided by the closed loop gain. So with unity gain, the common mode signal amplitude is the same as the output signal amplisiliconchip.com.au No driver transistors If you compare the amplifier circuits to our previously published amplifier designs such as the Ultra-LD Mk.3 or 20W Class-A Amplifier, you will find many similarities. As with the Ultra-LD Mk.3 amplifier, this design uses 2-pole frequency compensation. As a result, the headphone amplifier has particularly low distortion at high frequencies. For a detailed explanation of the advantages of 2-pole compensation, refer to the article published in the July 2011 issue on “Amplifier Compensation and Stability”. siliconchip.com.au Fig.7: total harmonic distortion and noise versus power with the larger filter capacitors and a 2A plugpack. Music power is 4.25W (both channels driven) but continuous output power is limited by the power supply. 1 THD+N vs Power, 1kHz, 8, 20Hz-22kHz Bandwidth 0.5 06/10/11 14:08:47 Both channels driven One channel driven Music power (both channels) 0.2 Total Harmonic Distortion + Noise (%) tude, which for our amplifier can be nearly 20V peak-to-peak. Typically, if the common mode signal exceeds 1-2V RMS, common mode distortion can become the dominant distortion mechanism, marring its performance. This is due to “Early effect” in the input transistors (named after James M. Early of Fairchild Semiconductor). This is caused by the effective width of the transistor base junction varying with its collector-base voltage (see http://en.wikipedia.org/wiki/ Early_effect). If the common mode voltage is large enough, the result is modulation of the input transistors’ beta and this reduces the overall linearity of the amplifier. These non-linearities cannot be corrected by negative feedback since they occur in the input stage. The solution is to use an inverting amplifier, as we have in this case. Its non-inverting input is connected to ground and so the inverting input is held at “virtual ground” too, regardless of the output voltage. This configuration has so little common mode voltage that it can’t suffer from common mode distortion. To make a power amplifier inverting, we rearrange the feedback network in the same manner as we would with an op amp. In fact, common mode distortion in op amps can be reduced using the same method. The main disadvantage of the inverting configuration is that the input impedance is low, as determined by the resistor from the signal source to the inverting input. For good noise performance, its value must be low (minimising its Johnson-Nyquist thermal noise). In this case, the preamplifiers provide the amplifiers with a low source impedance, so it isn’t a problem. 0.1 0.05 0.02 0.01 0.005 0.002 0.001 0.0005 0.0002 0.0001 50m 100m 200m 500m 1 2 Power (Watts) The main difference is that the two output transistors are driven directly from the voltage amplification stage, with no driver transistors in between. In this case, the output current is quite small due to the relatively low power, so we can get away without the driver stage as long as the output transistors have a good beta figure. In this case, we are using readily available TIP31 (NPN) and TIP32 (PNP) transistors, rated at 3A and 40W each; more than enough for our needs. They have an excellent beta for a power transistor, at around 200 for 100mA and 25°C. How it works Let’s start with the preamp stages and since both channels are identical, we will just describe the left channel. Any RF signals picked up by the input leads are attenuated by a low-pass filter consisting of a ferrite bead, a 100Ω resistor and a 100pF capacitor. The ferrite bead acts like an inductor to block RF. The signal is then coupled via a 470nF capacitor to pin 3 of op amp IC1a which is configured as a voltage follower. This provides a low source impedance to the preamp gain stages comprising IC2a & IC2b. IC1a’s output is fed to the following stage via a 220µF electrolytic capacitor. This large value ensures good bass response and avoids any distortion that may arise from the typical nonlinearity of an electrolytic capacitor. The signal passes to the non-inverting input of IC2a (pin 3) via volume control potentiometer VR1 and a 22µF electrolytic capacitor. This capacitor ensures there is no DC flowing through VR1, which would otherwise cause a crackling noise when it is rotated. IC2a buffers the voltage at the wiper of VR1 to provide a low impedance for inverting amplifier IC2b. IC2b has a fixed gain of 14.7, set by the 10kΩ and 680Ω resistors. The 100pF feedback capacitor is there to improve circuit stability and reduce high-frequency noise. Volume potentiometer VR1 is part of the feedback network from the output from IC2b to the input at the 220µF capacitor (from pin 1 of IC1a). Hence IC2a & IC2b form a feedback pair with the overall gain adjustable by VR1. When VR1 is rotated fully anticlockwise, IC2b’s output is connected directly to VR1b’s wiper. Thus IC2b is able to fully cancel the input signal (as there is zero impedance from its output to the wiper) and the result is silence (no output signal) from the preamplifier. Conversely, when VR1 is fully clockwise, VR1b’s wiper is connected directly to the input signal, which is then amplified by the maximum amount (14.7 times) by IC2b. At intermediate settings, the signal at the wiper is partially cancelled by the mixing of the non-inverted (input) and inverted (output) signals and the resulting gain is intermediate. The way in which this cancellation progresses as VR1 is varied provides a quasi-log law gain curve. IC1 needs input protection Because the headphone amplifier may be turned off when input signals September 2011  67 5 Parts List: Hifi Stereo Headphone Amplifier 1 PCB, code 01309111, 198 x 98mm 1 1U half rack case (Altronics H4995) (optional) 1 12V AC 1A or 2A plugpack 1 10kΩ dual gang linear 16mm potentiometer (VR1) 2 500Ω sealed horizontal trimpots (VR2, VR3) 1 PCB-mount white switched RCA socket (CON1) 1 PCB-mount red switched RCA socket (CON2) 1 PCB-mount DC socket (CON3) 1 PCB-mount 6.35mm stereo jack socket (3PST) with extended pins (Jaycar PS-0190 or equivalent) (CON4) 1 PCB-mount right-angle SPDT mini toggle switch (S1) (Altronics S1320) 2 M205 PCB-mount fuse clips (F1) 1 M205 1A fast-blow fuse (F1)* 6 PCB-mount 6021-type flag heatsinks (Element14 Order Code 1624531; Jaycar HH8504, Altronics H0637) 8 TO-220 insulating washers 6 TO-220 insulating bushes 2 plastic former bobbins (Jaycar LF1062, Altronics L5305) 1 2m length 0.8mm diameter enamelled copper wire 1 25mm length 25mm diameter heatshrink tubing 6 PCB pins 4 M3 x 15mm machine screws 6 M3 x 10mm machine screws 10 M3 nuts are present, IC1’s input transistors can be subjected to relatively high voltages; up to 2.5V RMS or maybe 7V peak-to-peak. This will not damage IC1 immediately but over many years, it could degrade the performance. This is because normally very little current flows through the op amp inputs and so the metal traces within the IC are thin. If enough current passes through the inputs (5mA or more), “metal migration” can cause degradation and ultimately failure. For that reason we have included small-signal Schottky diodes D15 & D16 to protect pin 3 of IC1a (and D17 & D18 for pin 5 of IC1b) when the unit is switched off but a large signal 68  Silicon Chip 18 M3 flat washers 4 M3 Nylon nuts with integral washers (Jaycar HP0150) or M3 Nylon nuts and washers 1 35 x 15mm section of tin plated steel (eg, cut from a tin can lid) 1 3mm black plastic LED clip (Jaycar HP1100, Altronics H1547) 1 knob to suit VR1 (suggested: Altronics H6213) 3 8-pin DIL sockets (optional) 2 small ferrite beads 1 250mm length 0.7mm diameter tinned copper wire Semiconductors 3 LM833 dual low noise op amps (IC1-IC3) 1 7812 positive 12V linear regulator (REG1) 1 7912 negative 12V linear regulator (REG2) 2 TIP31 3A NPN transistors (Q11, Q23) 2 TIP32 3A PNP transistors (Q12, Q24) 2 BD139 1.5A NPN transistors (Q10, Q22) 2 BC328 PNP transistors (Q25, Q26) 2 BC338 NPN transistors (Q9, Q21) 6 BC549 NPN transistors (Q3-Q4, Q8, Q15-Q16, Q20) 10 BC559 PNP transistors (Q1-Q2, Q5-Q7, Q13-Q14, Q17-Q19) 1 3mm blue LED (LED1) 14 1N4004 1A diodes (D1-14) is applied. They clamp the voltage at that input to within ±0.3V of the supply rails under normal conditions, preventing current flow through the op amp input transistors should their junctions be reverse-biased. So if the unit is off and the supply rails are zero, the input voltages will be similarly limited to ±0.3V. The BAT42 diodes have been carefully selected to clamp the op amp input voltages appropriately without having so much leakage current that they will introduce distortion into the signal (Schottky diodes normally have a much higher reverse leakage current than standard silicon diodes). For more information on protecting 4 BAT42 Schottky diodes (D15D18) (or use BAT85, Altronics Cat. Z0044) Capacitors 2 2200µF 25V electrolytic* 11 220µF 25V electrolytic** 4 47µF 16V electrolytic** 2 22µF 16V electrolytic** 2 470nF MKT 2 150nF MKT 7 100nF MKT 3 10nF MKT 2 680pF C0G/NP0 ceramic 2 220pF C0G/NP0 ceramic 4 100pF C0G/NP0 ceramic Resistors (0.25W, 1%) 4 100kΩ 2 680Ω 1 30kΩ 2 220Ω 3 22kΩ 6 100Ω 8 10kΩ 4 68Ω 10 2.2kΩ 2 47Ω 4 1.8kΩ 2 43Ω 2 1.1kΩ 4 22Ω 1 1kΩ 6 10Ω 2 910Ω 8 1.2Ω (1% or 5%) Notes * For driving speakers, upgrade the plugpack to 12V AC 2A, the fuse to 2A and the power supply capacitors to 4700µF 25V (diameter ≤16mm, height ≤30mm, eg, Futurlec C4700U25E105C). ** Low ESR 105° types can be used if their diameter is no more than 6.3mm for 22µF/47µF and 8mm for 220µF. op amp inputs, see Analog Devices tutorial MT-036, “Op Amp Output Phase-Reversal and Input Over-Voltage Protection”. We also tested BAT85 diodes (Altronics Z0044). These have slightly higher capacitance when reversebiased (10pF compared to 7pF) and a significantly higher reverse leakage current (400nA at -15V/25°C compared to 75nA). However, testing shows no measurable increase in distortion with these in place of the BAT42s so they are an acceptable substitute. Amplifier circuit Low-noise PNP transistors Q1 & Q2 are the differential input pair, with the siliconchip.com.au This view shows the fully assembled PCB. There’s no other wiring – you just assemble the board and install it in the case. base of Q1 being the non-inverting input to the amplifier and the base of Q2 being the inverting input. Q1’s base is tied to ground by a 910Ω resistor (to match the 900Ω source impedance at the base of Q2) and is bypassed by a 100nF capacitor to reduce highfrequency noise. The signal from the preamplifier is fed to the base of Q2 via a 1.8kΩ feedback resistor, so that the amplifier works in the inverting mode. 1.8kΩ is the lowest value resistance that IC2b can drive in parallel with its own feedback network. PNP transistor Q5 operates as a 3mA constant current source (0.65V ÷ 220Ω) to feed the Q1/Q2 input pair. Negative feedback for current regulation is provided by another PNP transistor, ie, Q6. It has a bootstrapped collector current sink (two 10kΩ resistors and a 47µF capacitor), so that it operates consistently. NPN transistors Q3 and Q4 form a current mirror for the input pair, with 68Ω emitter resistors to improve its accuracy. Any difference in the current through Q1 and Q2 must then flow to the base of NPN transistor Q8. So Q1Q5 form the transconductance stage of the amplifier. Together, Q8 and Q9 form a Darlington-like transistor, configured as a common-emitter amplifier. PNP transistor Q7 acts as a constant current source for its collector load, sourcing about 15mA (0.65V ÷ 43Ω). Q6 siliconchip.com.au provides current regulation feedback for Q7 as well as Q5. The 680pF and 220pF capacitors between Q9’s collector and Q8’s base, together with the 2.2kΩ resistor from their junction to the negative rail, form the 2-pole frequency compensation scheme mentioned earlier. Together, transistors Q7-Q9 are the voltage amplification stage. VBE multiplier Between Q7 and Q9 is Q10 which functions as a VBE multiplier to set the quiescent current for the output transistors Q11 & Q12. Q10 is mounted on the back of Q11’s heatsink so that its junction temperature tracks the output stage. Thus, its VBE tracks that of the output transistors (Q11 and Q12), so the bias voltage varies to compensate for changing output transistor temperature, keeping the standing current through them more or less constant. VR2 is used to adjust this current, while the 2.2kΩ resistor prevents the bias from becoming excessive if VR2’s wiper goes open circuit, as it may do while it is being trimmed. A 47µF capacitor filters the bias voltage, improving distortion performance when the output voltage swing is large. The resulting bias voltage is applied between the bases of output transistors Q11 (NPN) and Q12 (PNP) via 22Ω stopper resistors, which prevent parasitic oscillation. Each output transistor has a 0.6Ω emitter resistor (two 1.2Ω resistors in parallel) which helps to linearise the output stage and stabilise the quiescent current. Current limiting While it’s always a good idea to plug and unplug the headphones while the power switch is off, we can’t rely on that and we don’t want the output transistors to blow when it happens. Therefore, both Q11 and Q12 are protected against over-current conditions. Q11 is current-limited because the 15mA current source (Q7) sets a maximum limit for its base current. According to the TIP31 data sheet, at 25-125°C, the maximum collector current will be about 1.25A; well within its safe operating area (SOA) so as long as the short-circuit is brief. Q12 is more of a concern because Q9 can sink significantly more than 15mA. The 10kΩ resistor at Q8’s collector ultimately limits how much current Q9 can sink as follows. Q8’s maximum collector current is around (12V - 0.7V) ÷ (10kΩ + 2.2kΩ) = 1mA. Q9’s maximum current gain figure is around 165 (according to the BC338 data sheet), so the maximum Q9 can sink is about 165mA. Hence Q9 is a BC338 (a BC549 has a continuous collector current limit of 100mA). However, if this much current were sunk from Q12’s base then it would fully saturate (turn on hard), exceeding its SOA and possibly causing it to fail. September 2011  69 Silicon Chip Binders REAL VALUE AT $14.95 PLUS P & P These binders will protect your copies of S ILICON CHIP. They feature heavy-board covers & are made from a dis­ tinctive 2-tone green vinyl. They hold 12 issues & will look great on your bookshelf. H 80mm internal width H SILICON CHIP logo printed in gold-coloured lettering on spine & cover H Buy five and get them postage free! Price: $A14.95 plus $A10.00 p&p per order. Available only in Aust. Silicon Chip Publications PO Box 139 Collaroy Beach 2097 Or call (02) 9939 3295; or fax (02) 9939 2648 & quote your credit card number. Use this handy form Enclosed is my cheque/money order for $________ or please debit my  Visa    Mastercard Card No: _________________________________ Card Expiry Date ____/____ Signature ________________________ Name ____________________________ Address__________________________ __________________ P/code_______ 70  Silicon Chip Fig.8: the green trace in this scope grab shows the distortion residual for 100mW into 32Ω at 20kHz. Most of this is actually noise with very little harmonic content. Into lower load impedances (eg, 8Ω) the distortion becomes more apparent and is primarily third harmonic, with some higher harmonics. Q25 and D7 prevents this. Should the current flow through Q12’s collectoremitter junction exceed 2A (within its SOA), the drop across the 0.6Ω emitter resistor exceeds 2A x 0.6Ω = 1.2V. At this point, Q25’s base-emitter voltage increases beyond 1.2V - 0.6V = 0.6V and so Q25 starts to turn on, shunting current around Q12’s baseemitter junction and preventing Q12 from turning on harder. Any current sunk by Q9 beyond that necessary for Q12 to pass 2A goes through D7 and Q25 rather than Q12’s base-emitter junction. Output RLC filter The output filter isolates the amplifier from its load, improving stability. Because this amplifier circuit is already fairly stable (thanks to its simple output stage), we can get away with slightly less inductance than usual (4.7µH rather than 6.8µH or 10µH). We can thus use a thinner gauge wire which is slightly easier to wind, for roughly the same DC resistance. Ideally, the output filter should be optimised for the expected load impedance but because headphones have such a wide range of impedances, all we can do is compromise and specify an intermediate value. As a result, for higher impedances, the amplifier has a slightly elevated response at above 20kHz (see Fig.5). For 8Ω operation, there is a very slight roll-off at the high-frequency end of -0.02dB at 20kHz. At around 10-12Ω, the high frequency response will be virtually flat and then for higher load impedances, up to infinity, the gain is as much as +0.13dB at 20kHz. The increase is slightly lower (+0.09dB) for the most common impedances of 16Ω and 32Ω. This deviation is so small as to be imperceptible. In fact, all our amplifier designs using this type of output RLC filter (devised by Neville Thiele) have such a response with higher than usual output impedances or no load. Power supply The 12V AC plugpack plugs into an on-board DC connector (CON3). A 1A fuse protects the plugpack in case of a board fault or overload. The power switch (S1) is in the ground leg so that the tracks to and from it (near the edge of the PCB) have minimal AC voltage. This eliminates electrostatic radiation, preventing any coupling to nearby signal tracks. The incoming AC is half-wave rectified by diodes D1 & D2, with three siliconchip.com.au A half-size 1-unit steel case is used to house the assembled Headphone Amplifier PCB. Pt.2 next month has all the construction and setting-up details. 10nF metal film capacitors for RF and switching suppression. The resulting ±16V rails (nominal; under light load, closer to ±20V) are regulated to ±12V using 3-terminal regulators REG1 & REG2. So why are we regulating the supply for the whole device rather than just the op amps? Essentially it is because the amplifiers draw so little power when driving headphones that they might as well run off the regulated rails. In addition, the unregulated supply ripple is 50Hz because of the half-wave rectifiers (rather an 100Hz). The regulated supply rails give a lower hum and noise figure. Switch-on/off behaviour The circuit has been carefully designed to avoid loud thumps from the headphones when the unit is switched on or off. With a power amplifier, this is usually taken care of with an output muting relay that is also used for speaker protection. Since this amplifier has a low power output and a limited output current, a protection relay isn’t necessary. That is not say that you won’t hear any thumps at all. That will depend, in part, on the efficiency of your headphones. However, any thumps you do hear will be very slight. This has partly been achieved by removing the capacitor which would typically be between Q5’s base and siliconchip.com.au the positive rail (as present in the 20W Class-A Amplifier and the Ultra-LD Mk.3). This is not necessary with a regulated supply and if present, it delays the operation of the constant current source controlled by Q5 by several hundred milliseconds at switch-on. This would have caused a loud thump from the headphones had it been retained. Diodes D11 & D12 (D13 & D14 in the right channel) are also important for proper switch-on behaviour. While the ±12V regulated rails are already protected to prevent the positive rail from going negative and vice versa, the RC filtered supply rails for the early amplifier stages can still suffer from this problem unless extra steps are taken. That’s because the filter resistors isolate the capacitors from the clamp diodes D4 & D6. Without D11 and D12, the positive filtered rail could be briefly pulled negative and this would cause an amplifier output excursion. The different positive and negative rail filter resistors (10Ω and 47Ω respectively) allow the positive rail to come up more quickly which also helps achieve a clean switch-on. Together, these details allow the amplifiers to operate normally just milliseconds after both filter capacitors are partially charged. Similarly, diodes D9 & D10 clamp the RC-filtered supply for the op amps in the preamplifier. Without these, the op amp input transistors may become briefly reverse-biased at switch on, causing supply current to flow into the AC-coupling capacitors and again causing a thump to be generated. Finally, the 1kΩ resistor in parallel with D10 discharges the op amp negative supply rail faster than the positive rail when power is removed. The op amps are prone to oscillation when their supply capacitor is mostly discharged and this can cause a “chirp” at switch-off. With the 1kΩ discharge resistor, this chirp is made very short and often eliminated entirely. Increasing the output power While the circuit as presented is capable of driving loudspeakers, a few small changes can usefully increase the power output. If the 2200µF filter capacitors are changed to 4700µF, it increases the current they can supply before regulator drop-out begins. Also, a 12V AC 2A plugpack can be used in combination with a higher rated 2A fuse. This increases the available output power a little more. The THD+N vs power graph (Fig.7) shows the performance when both modifications are incorporated. Next month Next month, we shall present the construction details and describe the SC setting-up procedure. September 2011  71